Electronic device, electrostatic capacitance sensor and touch panel

ABSTRACT

An electronic apparatus includes a sensor system, an excitation generating unit which generates an intermittent sine wave signal and applies the same to the sensor system, and a demodulation unit which demodulates an amplitude modulated signal which is an output of the sensor system, in which the demodulation unit generates a demodulated signal using both a response of the sensor system in a period when the excitation generating unit outputs a sine wave and a response of the sensor system in a period, at least either immediately before or immediately after the above-mentioned period, when the excitation generating unit does not output a sine wave.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a Continuation of U.S. application Ser. No.14/434,982, filed Apr. 10, 2015, which is a National Stage ofInternational Application No. PCT/JP2013/077861, filed on Oct. 11, 2013,which claims priority from International Application No.PCT/JP2012/076370, filed on Oct. 12, 2012, the contents of all of whichare incorporated herein by reference in their entirety.

FIELD OF THE INVENTION

The present invention relates to an electronic device, a electrostaticcapacitance sensor and a touch panel and, in particular, to anelectronic device, a electrostatic capacitance sensor and a touch panelthat use an amplitude modulation and demodulation system.

BACKGROUND ART

A touch panel is a device that detects the coordinates of a positionpointed to by a pointing object such as a finger or a pen, or detects apointing action by such a pointing object. Today, touch panels aretypically used in combination with a display such as a Liquid CrystalDisplay (LCD), a Plasma Display Panel (PDP), or an organic EL display.

Outputs from a touch panel are input into a computer, which controlsimages displayed on the display and controls a device, therebyimplementing an easy-to-use human interface. Touch panels are now usedin a wide range of everyday-life applications such as video gamemachines, portable information terminals, ticket-vending machines,automatic teller machines (ATMs), and automobile navigation systems. Ascomputers grow more powerful and network connection environments becomewidespread, a wider variety of services are provided through electronicdevices and the need for displays with a touch panel is increasing.

One type of touch panel is a surface capacitive touch panel. The surfacecapacitive touch panel includes (i) a resistive sheet and (ii) drive andsensing circuitry which is connected to the resistive sheet, applies anAC voltage (sinusoidal voltage) to the resistive sheet as an excitation,measures a current flowing through the resistive sheet, and outputs themeasurement.

Specifically, the surface capacitive touch panel includes a transparentsubstrate, a transparent resistive sheet formed on the surface of thetransparent substrate, and a thin insulating film formed on the topsurface of the resistive sheet. The resistive sheet is referred to asthe position sensing conductive film. To drive the touch panel of thistype, an AC voltage is applied to the four corners of the positionsensing conductive film. When the touch panel is touched with a humanfinger, a pointing stick or the like (hereinafter referred to as theindicator), a capacitor is formed by capacitive coupling between theposition sensing conductive film and the indicator. A small currentflows to the indicator via the capacitor. The current flows from thecorners of the position sensing conductive film to the point touchedwith the indicator. A signal processing circuit calculates whether ornot there is a touch with an indicator and, the coordinate of theposition touched with the indicator on the basis of the current detectedby the drive and sensing circuitry. Specifically, the signal processingcircuit detects the presence or absence of a touch on the basis of thesum of currents from the four corners of the position sensing conductivefilm. In addition, the coordinates of the touch position is calculatedon the basis of the ratio of the currents from the four corners of theposition sensing conductive film.

Touch panels based on the surface capacitive operation principledescribed above are disclosed in Patent Literatures 1 to 5.

A device in Patent Literature 1, which includes a display panel and atouch panel combined and operated together, is equipped withcounter-electrode driving means for applying an AC voltage to the touchpanel during a non-display period of the display panel and applying thesame AC voltage to the counter electrodes of the display panel in orderto prevent decrease in the precision of position detection due to adrive signal for the display panel.

Patent Literature 2 discloses a “touch panel device in which when noiseis large, the level of AC voltage oscillation is increased whereas whennoise is small, the AC voltage oscillation level is decreased and, whenthere is noise with a specific frequency, switching is made to adifferent voltage oscillation frequency, thereby ensuring safety toachieve an improved signal-to-noise ratio, high noise robustness, andelectrical safety”.

Patent Literature 3 states that “a phase and an AC voltage when a fingerhas touched the panel are set as a contact vector signal and a scalarquantity calculated from the phase difference between the two signalsand amplitudes by using the cosine theorem is set as an AC signal of theactual finger touch, thereby excluding the AC voltage due to a parasiticsignal when a finger is not near the resistive sheet or the phasedifference between the signals due to a finger of a capacitive groundinghuman body or a resistive grounding human body from the detection of thetouch position”.

Patent Literature 4 discloses that “the operational circuit takes aninput of an output from the long sensor line LSLi and an output from theshort sensor line SSLi and performs a computation using the difference(Delta) between the outputs and the line capacitance ratio Kc to obtaina signal component S”.

Patent Literature 5 discloses that “the four nodes are labeled withsymbols Na, Nb, Nc and Nd. Terminals of current sensing circuits, whichwill be described later, are connected to the nodes” and states that“single-pole double-throw switches 21 a to 21 d are connected to thenodes Na to Nd through current sensing circuits 13 a to 13 d. An ACvoltage source 22 is connected to one of the two contacts of each of thesingle-pole double-throw switches 21 a to 21 d and a storage capacitanceline drive circuit is connected to the other contact (i.e. a nodelabeled with COM in FIG. 4). The waveform of an AC voltage may be asinusoidal wave, for example”.

On the other hand, the patent applicant including the present inventorpresented several proposals in Patent Document 6 and Non-Patent Document1 to attain an object of “detecting touching or non-touching, or a touchposition with high precision by removing noise whose frequency is thesame as a signal frequency or as close to the signal frequency as cannotbe resolved by a conventional frequency resolution”.

PRIOR ART LITERATURE Patent Literature

-   Patent Literature 1: Japanese Laid-open Patent Publication No.    2007-334606-   Patent Literature 2: Japanese Laid-open Patent Publication No.    2006-106853-   Patent Literature 3: Japanese Laid-open Patent Publication No.    2010-86285-   Patent Literature 4: Japanese Laid-open Patent Publication No.    2011-13757-   Patent Literature 5: Japanese Laid-open Patent Publication No.    2011-14109-   Patent Literature 6 Published Japanese Translation of PCT    International Publication for Patent Application No. 2011-069673

Non-Patent Document

-   Non-Patent Document 1 H. Haga et al., “A 10.4-in. On-Cell    Touch-Panel LCD with Correlated Noise Subtraction Method,” SID′ 12    Digest, pp 489-492 (2012) Society for Information Display 2012    International Symposium

SUMMARY OF THE INVENTION Problems to be Solved by the Invention

The following analysis has been made by the present inventors. The touchpanel described in Patent Literature 1 has the following six problems.

A first problem is that the touch panel is sensitive to external noise(electric field variations and capacitive coupling noise). While PatentLiterature 1 states that decrease in the precision of position detectiondue to the drive signal for the display panel is prevented, the touchpanel is susceptible to external noise from sources other than the drivesignal for the display panel, for example noise emitted from afluorescent lamp including an inverter circuit that is located above thetouch surface of the touch panel.

One reason for the problem is based on the operating principle of thetouch panel. Since a surface capacitive touch panel detects thecapacitance of a capacitor formed between the position sensingconductive film and a finger, a shied electrode for shielding anelectric field may not be formed between the position sensing conductivefilm and a finger. Therefore, the touch surface of the position sensingconductive film inevitably has a structure that is vulnerable toexternal noise. The larger the size of the touch panel, the moresusceptible to external noise the touch panel is.

Another reason is that the number of noise sources is increasing. Forexample, inverter fluorescent lamps developed for reducing flickeringare accepted in the marketplace and are increasing in number. Inaddition, more and more switching-mode power supplies, developed inorder to increase the efficiency of supply voltage conversion, are beingused in rechargers and AC adapters for portable devices. Noise generatedfrom these devices prevents normal operation of capacitance sensingdevices.

A second problem is that a bandpass filter may not remove noise when theexcitation frequency of the touch panel is equal or close to thefrequency of the noise.

The fundamental frequency of noise illustrated above or the frequency ofharmonic of noise is equal or close to the excitation frequency of thetouch panel. A synchronous detector described in Patent Literature 1 isclaimed to perform filtering in order to filter out noise withfrequencies different from the excitation frequency. Accordingly, themethod that decomposes an observed signal by frequency to select afrequency in this way may not remove noise that has the frequency equalto the excitation frequency.

When the frequency of noise is close to the excitation frequency, noisepasses through an attenuation band (or a transition band) between thepassband and the stopband of the bandpass filter hence the output of thebandpass filter contains noise. In other words, a practicable bandpassfilter has certain frequency resolution limits and therefore may notremove noise with frequencies that are close to the excitationfrequency.

A third problem is that if a touch sensing operation time period isrestricted by a non-display period (non-addressing period) or the like,frequency resolution decreases so that noise close to the frequency of atrue signal may not be removed. For example, when a signal of interestincludes two sinusoidal wave signals having the same amplitude, aperiodogram spectral estimation method can resolve such spectral peaksthat satisfy

$\begin{matrix}{{\Delta\; f} \geq \frac{1}{T}} & \left\lbrack {{Formula}\mspace{14mu} 1} \right\rbrack\end{matrix}$where T is a signal acquisition time period.

In the method, when the signal acquisition time period T is 500microseconds, Δf is 2 kHz, and therefore, when assuming a true signal of100 kHz and noise of 99 kHz, both may not be able to be resolved byfrequency.

A fourth problem is that the effect of noise removal by averagingdecreases and the signal-to-noise ratio decreases. For example, when anobserved signal on which Poisson distribution noise is superimposed isacquired many times and averaged to cancel out noise, thereby reducingthe noise, the amount of noise reduction is proportional to the squareroot of the number of the times the observed signal has been acquired.In other words, when the signal acquisition time period is limited to ashort time period such as non-display period (non-addressing period),the effect of the noise removal by averaging decreases and thesignal-to-noise ratio decreases.

A fifth problem is that if a structure in which a polarizer is placedbetween a position sensing conductive film and a finger is used asillustrated in Japanese Patent Application No. 2009-163401 by thepresent applicant, capacitance formed between the position sensingconductive film and the finger is reduced and the signal-to-noise ratiomay decrease. Similarly, if a protective glass or the like is insertedbetween the position sensing conductive film and a finger, thesignal-to-noise ratio may decrease.

A sixth problem, which is yet to be solved by the inventor and theapplicant of the present invention in Patent Document 6 and Non-PatentDocument 1, is a need for obtaining a high S/N ratio with respect tomore various noises. In Patent Document 6, |X_(n)−M_(n)| is calculatedfor every n (n is the number 1, 2, 3, . . . sequentially assigned to anon-address period) and a resultant value is taken as an output of ademodulating unit. The inventor of the present application, however,found that calculating |X_(n)−M_(n)| for every n might cause reductionin an S/N ratio in some cases.

Therefore, there is the problem of providing an electronic device, aelectrostatic capacitance sensor, and a touch panel that are capable ofremoving noise with a frequency equal to the frequency of a signal orclose to the frequency of a signal that the noise may not be resolvedwith conventional frequency resolutions and are therefore capable ofprecisely detecting the presence or absence of a touch and a touchposition.

Means for Solving the Problem

An electronic apparatus according to the present invention aiming atsolving the above problems is an electronic apparatus including a sensorsystem; an excitation generating unit which generates an intermittentsine wave signal and applies the same to the sensor system; and ademodulating unit which demodulates an amplitude modulated signal as anoutput of the sensor system, in which the demodulating unit generates ademodulated signal using both a response of the sensor system in aperiod when the excitation generating unit outputs a sine wave and aresponse of the sensor system in a period, at least either immediatelybefore or immediately after the aforementioned period, when theexcitation generating unit does not output a sine wave, and with avector as X, which is obtained from an amplitude and a phase of afrequency component of a sine wave that are calculated from the responseof the sensor system in a period when the excitation generating unitoutputs the sine wave, and with vectors as Y and Z, which are obtainedfrom an amplitude and a phase of a frequency component of a sine wavethat are respectively calculated from the response of the sensor systemin periods, immediately before and immediately after the aforementionedperiod, when the excitation generating unit does not output a sine wave,the demodulated signal corresponds to a constant multiplication of|X−k·M| in which M represents a mean vector of Y and Z, and k representsa coefficient whose value is determined using a response of the sensorsystem in a period when the excitation generating unit does not output asine wave.

In order to solve the above problems, an electrostatic capacitancesensor according to the present invention includes an electronicapparatus including a resistive sheet; and a sensor system configuredwith a driving and detecting circuit connected to the resistive sheetfor applying a voltage to the resistive sheet to measure and outputcurrent flowing through the resistive sheet, in which a touching stateor coordinates of an indicator are detected by detecting anelectrostatic capacitance of a capacitor formed by the resistive sheetand the indicator.

Or the electrostatic capacitance sensor includes an electrode; and asensor system configured with a driving and detecting circuit connectedto the electrode for applying a voltage to the electrode to measure andoutput current flowing through the electrode, in which a touching stateor coordinates of an indicator are detected by detecting anelectrostatic capacitance of a capacitor formed by the electrode and theindicator.

Further provided are a first electrode; a second electrode; and a sensorsystem configured with a driving circuit which applies a voltage to thefirst electrode and a detecting circuit which measures and outputscurrent flowing through the second electrode, in which a touching stateor coordinates of an indicator are detected by detecting anelectrostatic capacitance of a capacitor formed by the first electrodeand the second electrode.

A display device is further included, in which a non-address period ofthe display device has a period when the excitation generating unitoutputs a sine wave and a period when the unit does not output a sinewave, and a demodulated signal is generated using both a response of thesensor system in the period when a sine wave is output and a response ofthe sensor system in the period when the sine wave is not output.

In addition, in order to solve the above problems, a touch panelaccording to the present invention includes a resistive sheet; and asensor system configured with a driving and detecting circuit connectedto the resistive sheet for applying a voltage to the resistive sheet tomeasure and output current flowing through the resistive sheet, in whicha touching state or coordinates of an indicator are detected bydetecting an electrostatic capacitance of a capacitor formed by theresistive sheet and the indicator.

Or the touch panel includes an electrode; and a sensor system configuredwith a driving and detecting circuit connected to the electrode forapplying a voltage to the electrode to measure and output currentflowing through the electrode, in which a touching state or coordinatesof an indicator are detected by detecting an electrostatic capacitanceof a capacitor formed by the electrode and the indicator.

On the other hand, in order to solve the above problems, the touch panelaccording to the present invention includes a first electrode; a secondelectrode; and a sensor system configured with a driving circuit whichapplies a voltage to the first electrode and a detecting circuit whichmeasures and outputs current flowing through the second electrode, inwhich a touching state or coordinates of an indicator are detected bydetecting an electrostatic capacitance of a capacitor formed by thefirst electrode and the second electrode.

Besides, in order to solve the above problems, the electronic apparatusaccording to the present invention is configured to include a displaydevice, in which a non-address period of the display device has a periodwhen the excitation generating unit outputs a sine wave and a periodwhen the unit does not output a sine wave, so that a demodulated signalis generated using both a response of the sensor system in the periodwhen the sine wave is output and a response of the sensor system in theperiod when the sine wave is not output.

By contrast, the touch panel is configured to include a display device,in which a non-address period of the display device has a period whenthe excitation generating unit outputs a sine wave and a period when theunit does not output a sine wave, so that a demodulated signal isgenerated using both a response of the sensor system in the period whenthe sine wave is output and a response of the sensor system in theperiod when the sine wave is not output.

It is noted that although the present specification and claims recitethat the excitation generating unit outputs a sine wave, the output inthis case is not limited to a sine wave having a single frequency. Everysignal can be represented as a series of sine waves having differentfrequencies (a Fourier series expansion). In other words, when theexcitation generating unit outputs a square wave, for example, thesquare wave expresses a series of sine waves having differentfrequencies. In this case, a demodulated signal may be obtained bysignal processing, focusing on a sine wave having a fundamentalfrequency of the square wave. Thus, the present invention includes evena case where the excitation generating unit outputs a square wave. Fromthe similar reason, the present invention includes any case where theexcitation generating unit outputs any alternating current.

Effects of the Invention

By implementing an electronic device, an electrostatic capacitancesensor, a touch sensor and a touch panel according to the presentinvention, the following six effects can be obtained.

A first effect is that since noise is acquired by stopping a sinusoidalwave, the noise can be accurately acquired regardless of the presence orabsence of a finger (presence or absence of a touch).

A second effect is that since the signal processing path for “noise”acquired by stopping a sinusoidal wave is the same as the signalprocessing path for “true signal plus noise” acquired by providing thesinusoidal wave, the noise can be accurately acquired.

A third effect is that since a subtraction is performed between vectorsof “true signal plus noise” and “noise”, the true signal can beaccurately obtained even when the true signal and the noise have thesame frequency.

A fourth effect is that noise with frequencies close to the frequency ofa true signal can be removed beyond frequency resolution by using themean vector of forward noise (noise acquired during a stop of asinusoidal wave before the excitation generating unit outputs thesinusoidal wave) and backward noise (noise acquired during a stop of thesinusoidal wave after the excitation generating unit has output thesinusoidal wave).

A fifth effect is that by using the mean vector of forward noise andbackward noise, noise can be precisely removed even when the amplitudeof the noise has changed.

A sixth effect is enabling a high S/N ratio to be obtained for variousnoises. In particular, applying a control means according to the presentinvention enables the problem of S/N ratio reduction to be avoided in anenvironment where an inverter circuit stops, while enabling an increasein an S/N ratio in an environment where the inverter circuit operates.

Owing to the six effects described above, the present invention enablesprovision of a touch panel and an electronic device that are robust toexternal noise and have a high signal-to-noise ratio.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of an electronic apparatus in ExemplaryEmbodiment 1 of the present invention;

FIG. 2 is a block diagram of an electrostatic capacitance sensor inExemplary Embodiment 1 of the present invention;

FIG. 3 is a timing chart of the electrostatic capacitance sensor inExemplary Embodiment 1 of the present invention;

FIG. 4 is a graph of a sensor system output voltage for explainingoperation of the present invention;

FIG. 5 is a vector diagram illustrating operation of a demodulating unitin Exemplary Embodiment 1 of the present invention;

FIG. 6 is a diagram illustrating a demodulating unit input voltage forexplaining operation in Exemplary Embodiment 2 of the present invention;

FIG. 7 is a vector diagram for explaining a demodulating unit inExemplary Embodiment 2 of the present invention;

FIG. 8 is a vector diagram for explaining the demodulating unit inExemplary Embodiment 2 of the present invention;

FIG. 9 is a vector diagram for explaining the demodulating unit inExemplary Embodiment 2 of the present invention;

FIG. 10 is a block diagram of an electrostatic capacitance sensor inExemplary Embodiment 3 of the present invention;

FIG. 11 is a diagram illustrating a configuration of an electrostaticcapacitive touch panel in Example 1 of the present invention;

FIG. 12 is a timing chart of the electrostatic capacitive touch panel inExample 1 of the present invention;

FIG. 13 is a diagram illustrating a signal waveform of the electrostaticcapacitive touch panel in Example 1 of the present invention;

FIG. 14 is a block diagram for explaining signal processing of theelectrostatic capacitive touch panel in Example 1 of the presentinvention;

FIG. 15 is a diagram illustrating a signal waveform of the electrostaticcapacitive touch panel in Example 1 of the present invention;

FIG. 16 is a flow chart of processing executed in a demodulating unit inExample 2 of the present invention;

FIG. 17 is a phase graph obtained in the demodulating unit when aninverter circuit is turned ON, in which (a) represents an angle[Y_(1,s)] and (b) represents an angle [Z_(1,s)] in Example 2 of thepresent invention;

FIG. 18 is a phase graph obtained in the demodulating unit when theinverter circuit is turned OFF, in which (a) represents an angle[Y_(1,s)] and (b) represents an angle [Z_(1,s)] in Example 2 of thepresent invention;

FIG. 19 is a phase graph obtained in the demodulating unit when theinverter circuit is turned ON by estimating a direction and an amount ofphase rotation with respect to the graph in FIG. 17, in which (a)represents an angle [Y_(1,s)] and (b) represents an angle [Z_(1,s)] inExample 2 of the present invention;

FIG. 20 is a phase graph obtained in the demodulating unit when theinverter circuit is turned OFF by estimating a direction and an amountof phase rotation with respect to the graph in FIG. 18, in which (a)represents an angle [Y₁,s] and (b) represents an angle [Z₁,s] in Example2 of the present invention;

FIG. 21 is a graph illustrating an S/N ratio of an electrostaticcapacitive touch panel when the inverter circuit is turned ON in Example2 of the present invention;

FIG. 22 is a graph illustrating an S/N ratio of the electrostaticcapacitive touch panel when the inverter circuit is turned OFF inExample 2 of the present invention;

FIG. 23 is a flow chart of processing of a demodulating unit in Example3 of the present invention;

FIG. 24 is a graph illustrating an S/N ratio of an electrostaticcapacitive touch panel when an inverter circuit is turned ON in Example3 of the present invention;

FIG. 25 is a graph illustrating an S/N ratio of the electrostaticcapacitive touch panel when the inverter circuit is turned OFF inExample 3 of the present invention;

FIG. 26 is a flow chart of processing of a demodulating unit in Example4 of the present invention;

FIG. 27 is a diagram illustrating a configuration of an electrostaticcapacitive touch panel in Example 5 of the present invention;

FIG. 28 is a timing chart of the electrostatic capacitive touch panel inExample 5 of the present invention;

FIG. 29 is a block diagram of an electrostatic capacitive touch panel inExample 6 of the present invention; and

FIG. 30 is a timing chart of the electrostatic capacitive touch panel inExample 6 of the present invention.

MODES FOR CARRYING OUT THE INVENTION Embodiment 1

A electrostatic capacitance sensor of the present invention will beillustrated. The function of a typical electrostatic capacitance sensoris implemented by omitting the function of detecting a position from thefunctions of a touch panel illustrated in the background art section.Since the position detection function is omitted, a conductive sheet orsimply a conductor may be used in place of the resistive sheet.

(Configuration)

FIG. 2 is a block diagram of a electrostatic capacitance sensor 100 ofthe present invention and FIG. 1 is a block diagram of an electronicdevice 120 of the present invention which is an abstraction of theelectrostatic capacitance sensor of the present invention. Theelectrostatic capacitance sensor 100 illustrated in FIG. 2 is configuredto detect the capacitance of a capacitor C_(in) depicted in the figure.The electrostatic capacitance sensor includes a sensor system 101 thattakes inputs of the capacitance of the capacitor C_(in) and anexcitation and outputs a signal according to the capacitance of thecapacitor C_(in), an excitation generating unit 102 generating theexcitation, a sinusoidal wave generating unit 103 connected to theexcitation generating unit, and a DC generating unit 104. The outputfrom the sensor system is input into a demodulating unit 105, whichgenerates a demodulated signal.

The excitation generating unit generates an intermittent sinusoidal wavesignal. Means for generating an intermittent sinusoidal wave signalincludes the sinusoidal wave generating unit 103 and the DC generatingunit 104 provided as illustrated in FIG. 2, and means for switchingtherebetween is provided. However, it is not limited to such means.Other means may be, for example, to use a digital-to-analog converterand provide a digital signal obtained by discretizing an intermittentsinusoidal wave to the DA converter.

The sensor system includes an operational amplifier 110, a resistanceR_(f) inserted in its feedback path, and a capacitor C_(f) and furtherincludes an adder 111 that performs a subtraction between an outputvoltage from the operational amplifier 110 and an excitation voltage.

Assuming that the operational amplifier 110 is an ideal operationalamplifier and letting V₁ denote the voltage of excitation input into thesensor system 101 and V₂ denote the output voltage from the sensorsystem, then the frequency response H(jω) of the sensor system can beobtained by solving a circuit equation obtained from the figure asfollows:

$\begin{matrix}{{H\left( {j\;\omega} \right)} = {\frac{V_{2}}{V_{1}} = \frac{j\;\omega\; C_{in}R_{f}}{1 + {j\;\omega\; C_{f}R_{f}}}}} & \left\lbrack {{Formula}\mspace{14mu} 2} \right\rbrack\end{matrix}$Here, ω represents the angular frequency of the excitation and jrepresents an imaginary unit.From the equation, the amplitude response |H(jω)| of the sensor systemis

$\begin{matrix}{{{H\left( {j\;\omega} \right)}} = \frac{\omega\; C_{in}R_{f}}{\sqrt{1 + {\omega^{2}C_{f}^{2}R_{f}^{2}}}}} & \left\lbrack {{Formula}\mspace{14mu} 3} \right\rbrack\end{matrix}$

As represented by formula 3, the amplitude of the output from the sensorsystem 101 is proportional to the capacitance of the capacitor C_(in).

Since the frequency of the output from the sensor system is equal to thefrequency of the excitation and the amplitude of the output changes inaccordance with the capacitance of the capacitor C_(in), the sensorsystem can be said to be an amplitude modulation system.

FIG. 2 can be abstracted to FIG. 1. The input S(t) into the sensorsystem can be a capacitance as illustrated in this embodiment as well asan electrical signal such as a voltage or a current.

(Operation)

An operation of the electrostatic capacitance sensor of the presentinvention will be illustrated with reference to FIG. 3.

The excitation generating unit 102 generates an intermittent sinusoidalvoltage as illustrated as the waveform at the top of FIG. 3, i.e.excitation generating unit output voltage. The sinusoidal voltage isprovided to the sensor system 101 as an excitation. The frequency of thesinusoidal wave in this example is 100 kHz. In response to theexcitation and the capacitance of the capacitor C_(in), the sensorsystem outputs a voltage f(t) as illustrated as the second waveform inFIG. 3, i.e. a sensor system output voltage. Responses of the sensorsystem in periods in which the excitation generating unit 102 isoutputting the sinusoidal wave are denoted as x₁(t), x₂(t) as in thefigure and output voltages from the sensor system in periods in whichthe excitation generating unit stops outputting the sinusoidal wave aredenoted as z₁(t), z₂(t).

According to formula 2, the amplitude of the output voltage from thesensor system is zero in the periods in which the excitation generatingunit stops outputting the wave. In reality, however, the amplitude isnot zero because of noise contamination. In many applications such astouch sensors and touch panels, the capacitance of the capacitor C_(in)as illustrated in FIG. 2 is the capacitance of a capacitor formed by anindicator and a resistive sheet, and external noise (electric fieldvariations and capacitive coupling noise) is easily coupled into theresistive sheet that constitutes a part of the capacitor C_(in). Thereason why z₁(t) and z₂(t) in FIG. 3 are not zero is that they reflectthe influence of the noise. When external noise is steady, there isexternal noise contamination regardless of whether there is sinusoidalwave excitation or the sinusoidal wave is stopped (DC), therefore noiseis present in x₁(t) and x₂(t). In other words, a true signal plus noise(true signal+noise) appears in x₁(t) and x₂(t) and only noise appears inz₁(t) and z₂(t).

A significant finding by the inventors is that z₁(t) and z₂(t) are notdependent on the capacitance of the capacitor C_(in) but representsexternal noise. In other words, in a touch sensor or a touch panel, onlynoise appears regardless of the presence or absence of a finger, whichis an indicator. This is because the impedance of the capacitor C_(in)formed by the finger and the position sensing conductive film issufficiently higher than the impedance of the sensor system, noiseentering the position sensing conductive film is coupled into the sensorsystem as a current regardless of the presence or absence of a finger.

Another significant finding is that there is a correlation between noisepresent in the sensor system output voltage in a period during which theexcitation generating unit is outputting the sinusoidal wave and noisepresent in the sensor system output voltages in the periods precedingand succeeding that period.

The demodulating unit 105 receives an output signal from the sensorsystem 101 and takes advantage of the features illustrated above toremove noise. An example will be illustrated where a true signal inx₁(t), here the amplitude of a true signal in x₁(t), is obtained from anobserved signal x₁(t) including the true signal plus noise and anobserved signal z₁(t) including only noise.

The demodulating unit 105 periodically reads a signal value from thesensor system output voltage f(t) at time intervals Δt and converts thesignal value into a discrete time signal f(iΔt), where i∈Z (Z is a setof integers). By sampling x₁(t) in this way, x₁(iΔt) is obtained, wherei=0, 1, 2, . . . , N−1, and by sampling z₁(t)), z₁(iΔt) is obtained,where i=0, 1, 2, . . . , Q−1.

Let X₁ denote the discrete Fourier transform Dk that corresponds to 100kHz which is the frequency of the excitation sinusoidal wave among thediscrete Fourier transforms Dk of x₁(iΔt), then a complex number X₁ canbe obtained as

$\begin{matrix}{X_{1} = {\frac{1}{N}{\sum\limits_{i = 0}^{N - 1}{{x\left( {i\;\Delta\; t} \right)}{\exp\left( {{- j}\; 2\pi\; 100\mspace{11mu}{kHz}\mspace{11mu} i\;\Delta\; t} \right)}}}}} & \left\lbrack {{Formula}\mspace{14mu} 4} \right\rbrack\end{matrix}$where j is an imaginary unit and N is the number of samples. The complexnumber X₁ can be written as a two-dimensional vector X₁≡(Re {X₁}, Im{X₁}), where Re {X₁} is the real part of the complex number X₁ and Im{X₁} is the imaginary part of the complex X₁. These are equivalent toeach other.

Similarly, let Z₁ denote the discrete Fourier transform Dk thatcorresponds to 100 kHz which is the frequency of the sinusoidal waveamong the discrete Fourier transforms Dk of z₁(iΔt), then a complexnumber Z₁ can be obtained as

$\begin{matrix}{Z_{1} = {\frac{1}{Q}{\sum\limits_{i = 0}^{Q - 1}{{z\left( {i\;\Delta\; t} \right)}{\exp\left( {{- j}\; 2\;\pi\; 100\mspace{11mu}{kHz}\mspace{11mu} i\;\Delta\; t} \right)}}}}} & \left\lbrack {{Formula}\mspace{14mu} 5} \right\rbrack\end{matrix}$where j is an imaginary unit and Q is the number of samples. The complexZ₁ can be written as a two-dimensional vector Z₁≡(Re {Z₁}, Im {Z₁}).These are equivalent to each other.

Assuming that the 100-kHz component of noise present in the observedsignal x₁(t) is the same as the 100-kHz component of the observed signalz₁(t), vector X₁−vector Z₁ is then calculated. The magnitude |X₁−Z₁| ofthe result is set as the amplitude of the true signal of x₁(t) and as ademodulated signal D(t) output from the demodulating unit.

The operation of the demodulating unit illustrated above will beillustrated by using a model of an observed signal and assigningspecific numerical values.

The model of the observed signal is illustrated in FIG. 4. Let f(t)denote the model of the observed signal, then f(t) is true signal(V_(sig)) with an amplitude of 2 V plus noise (V_(noise)) with anamplitude of 1 V, as follows:

$\begin{matrix}{{f(t)} = {{Vsig} + {Vnoise}}} & \left\lbrack {{Formula}\mspace{14mu} 6} \right\rbrack \\{{Vsig} = \left\{ \begin{matrix}{2\;{\sin\left( {2\pi\; 100\mspace{11mu}{kt}} \right)}} & \left( {{0.1\mspace{11mu} m{\;\;}\sec} < t < {0.3\; m{\;\;}\sec}} \right) \\0 & ({else})\end{matrix} \right.} & \left\lbrack {{Formula}\mspace{14mu} 7} \right\rbrack \\{{Vnoise} = {\sin\left( {{2\pi\mspace{11mu} 100\;{kt}} + {\frac{3}{4}\pi}} \right)}} & \left\lbrack {{Formula}\mspace{14mu} 8} \right\rbrack\end{matrix}$Sampling was performed at intervals of Δt=0.1 microseconds to discretizef(t) to f(aΔt), where a=0, 1, 2, . . . , 4999.

x₁(iΔt) and z₁(iΔt) are signals illustrated in FIG. 4. Considering thata 100-kHz component is to be extracted later, it is desirable that thelength (time) of x₁(iΔt), i.e. t₁′−t₁ be set to an integer multiple ofintervals of 100 kHz, i.e. n×10 microseconds, where n is a positiveinteger.

Specifically, x₁(iΔt), where i=0 to 1999, was set as f(aΔt), wherea=1000 to 2999, and t₁′−t₁ was set to 200 microseconds (n=20).

It is desirable that the beginning time t₂ of z₁(iΔt) be set to t₁+m×10μsec, where m is a positive integer. It is desirable that the length(time) of z₁(t), i.e. t₂′−t₂ be set to an integer multiple of intervalsof 100 kHz, i.e. w×10 microseconds, where w is a positive integer.

Specifically, z₁(iΔt), where i=0 to 1999, was set as f(aΔt), wherea=3000 to 4999, t₂=t₁+200 microseconds (m=20) was set, and t₂′−t₂ wasset to 200 microseconds (w=20).

X₁ and Z₁ were calculated to obtain the following results.

$\begin{matrix}{X_{1} = {{\frac{1}{2000}{\sum\limits_{i = 0}^{1999}\;{{x\left( {i\;\Delta\; t} \right)}{\exp\left( {{- j}\; 2{\pi 100}\;{kHz}\mspace{11mu} i\;\Delta\; t} \right)}}}} = {0.3536 = {j\; 0.6464}}}} & \left\lbrack {{Formula}\mspace{14mu} 9} \right\rbrack \\{Z_{1} = {{\frac{1}{2000}{\sum\limits_{i = 0}^{1999}\;{{z\left( {i\;\Delta\; t} \right)}{\exp\left( {{- j}\; 2{\pi 100}\;{kHz}\mspace{11mu} i\;\Delta\; t} \right)}}}} = {0.3536 = {j\; 0.3536}}}} & \left\lbrack {{Formula}\mspace{14mu} 10} \right\rbrack\end{matrix}$

The complex numbers given above are considered to be vectors, and vectorX₁, vector Z₁, and vector X₁−vector Z₁ are plotted on a complex plane asin FIG. 5.

The magnitude of vector X₁−vector Z₁ is 1.0 as in the figure. Byfocusing attention on the fact that the magnitude of each vector in FIG.5 is ½ of the amplitude of a signal of 100 kHz, the amplitude of thetrue signal was calculated as vector X₁−vector Z₁ equals 2 V. On theother hand, it is difficult to derive the amplitude (2 V) of the truesignal based only on information such as the amplitude 2×|X₁|(1.5 V) ofcalculated x₁(iΔt) and the amplitude 2×|Z₁|(1.0 V) of calculatedz₁(iΔt).

The amplitude (1.5 V) of x₁(iΔt) and the amplitude (1.0 V) of z₁(iΔt)are equivalent to calculated amplitudes of 100-kHz components of signalsx₁(iΔt) and z₁(iΔt), respectively. In other words, conventional noiseremoval using frequency separation alone may not obtain the amplitude ofthe true signal.

In the foregoing, an example has been given in which X₁ and Z₁ arecalculated from x₁(iΔt) and z₁(iΔt) and |X₁−Z₁| is calculated to obtainone value of the demodulated signal D(t). For the next value of D(t), X₂and Z₂ are calculated from x₂(t) and z₂(t) and |X₂−Z₂| is calculated asillustrated in FIG. 3. For the subsequent values of D(t), calculationsare performed in a similar manner to obtain a demodulated signal D(t).

The embodiment has two effects. The first effect is that since noise isacquired while a sinusoidal wave is stopped, noise can be accuratelyacquired regardless of the presence or absence of a finger or even whena finger has been placed on or removed from the panel or the pressureapplied by a finger has varied to change the capacitance of thecapacitor C_(in).

The second effect is that since a subtraction is performed betweenvectors of “true signal plus noise” and “noise”, the true signal can beaccurately obtained even when the true signal and the noise have thesame frequency.

Embodiment 2

In the embodiment 1, an observed signal z₁(iΔt) was used to obtain theamplitude of the true signal of an observed signal x₁(iΔt). In otherwords, noise z₁(iΔt) observed after an observed signal x₁(iΔt) was usedto remove noise. In the embodiment 2, a mode in which noise before andafter an observed signal x₁(iΔt) is used to obtain the amplitude of atrue signal of the observed signal x₁(iΔt) will be illustrated with thefocus on an operation of a demodulating unit.

FIG. 6 illustrates a model f(aΔt) of an observed signal obtained bydiscretizing a signal input in the demodulating unit 105, where a=0, 1,2, . . . , and Δt=0.4 microseconds.

f(aΔt) is a true signal (V_(sig)) with an amplitude of 1 V plus noise(V_(noise)) of 99 kHz whose amplitude changes with time. This can bemathematically written as follows:

$\begin{matrix}{{f\left( {a\;\Delta\; t} \right)} = {{Vsig} + {Vnoise}}} & \left\lbrack {{Formula}\mspace{14mu} 11} \right\rbrack \\{{Vsig} = \left\{ \begin{matrix}{\;{\sin\left( {2\pi\; 100\mspace{11mu} k\mspace{11mu} a\;\Delta\; t} \right)}} & \left( {4229 < a < 5879} \right) \\0 & ({else})\end{matrix} \right.} & \left\lbrack {{Formula}\mspace{14mu} 12} \right\rbrack \\{{Vnoise} = {\frac{a\;\Delta\; t}{4 \times 10^{- 3}}{\sin\left( {{2\pi\; 99k{\;\;}a\;\Delta\; t} + \pi} \right)}}} & \left\lbrack {{Formula}\mspace{14mu} 13} \right\rbrack\end{matrix}$where y(iΔt), x(iΔt) and z(iΔt) are signals extracted, respectively,from f(aΔt) as follows.

y(iΔt), where i=0 to 399 was set as f(aΔt), where a=3800 to 4199;x(iΔt), where i=0 to 1624, was set as f(aΔt), where a=4250 to 5874; andz(iΔt), where i=0 to 299, was set as f(aΔt), where a=6000 to 6299.

For convenience, y(iΔt) is herein referred to as forward noise andz(iΔt) is referred to as backward noise.

In the demodulating unit, the same method as that in the embodiment 1 isused to obtain complex numbers Y_(m) and Z_(m) from observed signalsy(iΔt) and z(iΔt) according to the following formulas.

$\begin{matrix}{Y_{m} = {\frac{1}{400}{\sum\limits_{i = 0}^{399}{{y\left( {i\;\Delta\; t} \right)}{\exp\left( {{- j}\; 2\pi\; 100\mspace{11mu}{kHz}\mspace{11mu} i\;\Delta\; t} \right)}}}}} & \left\lbrack {{Formula}\mspace{14mu} 14} \right\rbrack \\{Z_{m} = {\frac{1}{300}{\sum\limits_{i = 0}^{299}{{z\left( {i\;\Delta\; t} \right)}{\exp\left( {{- j}\; 2\pi\; 100\mspace{11mu}{kHz}\mspace{11mu} i\;\Delta\; t} \right)}}}}} & \left\lbrack {{Formula}\mspace{14mu} 15} \right\rbrack\end{matrix}$

Here, Δt is the sampling interval and j is an imaginary unit.

The vectors Y_(m) and Z_(m) obtained here are schematically illustratedin FIG. 7.

From the vectors Y_(m) and Z_(m), noise vectors Y and Z at time instantst₁ and t_(1′) are then estimated. The estimation method is as follows.Let Y_(m) be the noise vector at time instant (t₀+t_(0′))/2 and Z_(m) bethe noise vector at time instant (t₂+t_(2′))/2.

Approximation is made that the amplitudes and phases of the vectorschange from Y_(m) to Z_(m) with time, and the noise vectors Y and Z attime instants t₁ and t_(1′) are obtained. FIG. 7 schematicallyillustrates the relationship between Y_(m), Z_(m) and Y, Z.

Then, from the vectors Y and Z, the mean vector M of the vectors Y and Zis calculated. The calculation of the mean vector will be illustratedwith reference to FIG. 8.

As mentioned earlier, the vector representation and the complexrepresentation are equivalent to each other. The formula for calculatingM can be written in complex representation as follows:

$\begin{matrix}{M = {{\frac{1}{T}{\int_{0}^{T}{\left( {A_{S} - {\frac{A_{S} - A_{E}}{T}t}} \right){\cos\left( {\theta_{S} - {\frac{\theta_{S} - \theta_{E}}{T}t}} \right)}d\; t}}} + {j\left\{ {\frac{1}{T}{\int_{0}^{T}{\left( {A_{S} - {\frac{A_{S} - A_{E}}{T}t}} \right){\sin\left( {\theta_{S} - {\frac{\theta_{S} - \theta_{E}}{T}t}} \right)}d\; t}}} \right\}}}} & \left\lbrack {{Formula}\mspace{14mu} 16} \right\rbrack\end{matrix}$Here, T represents t_(1′)−t₁ in FIG. 6, A_(S) and θ_(S) represent theamplitude and phase of the vector Y, respectively, and A_(E) and θ_(E)represent the amplitude and phase of the vector Z, respectively. FIG.9(a) illustrates Y, Z and M obtained from the model signal in FIG. 6according to the foregoing.

Then, X is obtained in the same way as in the embodiment 1 and X−M iscalculated. X can be written as follows:

$\begin{matrix}{X = {\frac{1}{1625}{\sum\limits_{i = 0}^{624}{{x\left( {i\;\Delta\; t} \right)}{\exp\left( {{- j}\; 2\pi\; 100\mspace{11mu}{kHz}\mspace{11mu} i\;\Delta\; t} \right)}}}}} & \left\lbrack {{Formula}\mspace{14mu} 17} \right\rbrack\end{matrix}$

Here, Δt is the sampling interval and j is an imaginary unit.

FIG. 9(b) illustrates X obtained from x(iΔt) in FIG. 6, and M and X−Mobtained earlier.

From FIG. 9(b), |X−M| is 0.5, and taking note of the fact that thisvalue is ½ of the amplitude of the true signal, it has been confirmedthat the amplitude of the true signal, 1.0 V can be correctly obtained.In other words, it has been shown that if noise of 99 kHz, which is veryclose to the excitation frequency, 100 kHz, is present, the noise can beprecisely removed.

Generally, when the signal acquisition periods are limited as in thecase of x(iΔt), frequency resolution is reduced and noise close to thefrequency of a true signal may not be removed. In this embodiment, onthe other hand, noise close to the frequency of a true signal can beremoved beyond frequency resolution by using noise y(iΔt) preceding tox(iΔt) and noise z(iΔt) succeeding to x(iΔt) as illustrated.

Further, as illustrated in this embodiment, the mean vector M can beused to precisely remove noise even when the amplitude of the noise isdependent on time.

(Effects)

The embodiment has the following two effects.

First, by using the mean vector calculated from forward noise andbackward noise, noise close to the frequency of a true signal can beremoved beyond frequency resolution.

Second, by using the mean vector calculated from forward noise andbackward noise, noise can be precisely removed even when the amplitudeof noise has changed.

Embodiment 3

Embodiment 3 is an embodiment of application of the present invention toan electrostatic capacitance sensor.

(Structure)

FIG. 10 illustrates a block diagram of an electrostatic capacitancesensor 206 of the present invention. The electrostatic capacitancesensor 206 illustrated in FIG. 10 is configured to detect anelectrostatic capacitance of the capacitor C_(in) illustrated in FIG.10. The electrostatic capacitance sensor 206 has a sensor system 205which outputs a signal according to the electrostatic capacitance of thecapacitor C_(in), with the electrostatic capacitance of the capacitorC_(in) and an excitation as inputs, the excitation generating unit 102which generates the excitation, the sine wave generating unit 103connected to the excitation generating unit 102, and the DC generatingunit 104. An output of the sensor system 205 is input to thedemodulating unit 105, so that the demodulating unit 105 generates ademodulated signal.

The above-described electrostatic capacitance sensor in FIG. 2 isconfigured to detect an electrostatic capacitance of the capacitorC_(in) connected to the ground. By contrast, the electrostaticcapacitance sensor 206 in FIG. 10 as the present exemplary embodimentdetects an electrostatic capacitance of the capacitor C_(in) connectedbetween a node N1 and a node N2. In addition, the electrostaticcapacitance sensor in FIG. 10 is capable of detecting an electrostaticcapacitance of the capacitor C_(in) without being affected even when aparasitic capacitance is formed between the node N1 and the ground.

The excitation generating unit 102 generates an intermittent sine wavesignal.

The sensor system 205 is configured with a current voltage converter(I-V converter) 207, one electrode (electrode connected to the node N1)configuring the capacitor C_(in), and a DC bias circuit 208 (or a groundonly) connected to the I-V converter 207.

The I-V converter 207 is configured with the operational amplifier 110,a resistor R_(f) inserted in a feedback path, and the capacitor C_(f)inserted in the feedback path, in which other electrode (electrodeconnected to the node N2) configuring the capacitor C_(in) is connectedto an inverting input terminal of the operational amplifier 110.

Assuming the operational amplifier 110 as an ideal operationalamplifier, and setting a voltage of an excitation input to the sensorsystem 205 as V₁ and an output voltage of the sensor system 205 as V₂,the frequency response H(jω) of the sensor system 205 becomes the sameas that represented by [Formula 2] by solving a circuit formula obtainedfrom FIG. 10. Then, the amplitude response |H(jω)| of the sensor systembecomes the same as that represented by [Formula 3]. An output amplitudeof the sensor system 205 is accordingly proportional to an electrostaticcapacitance of the capacitor C_(in).

In addition, since the output of the sensor system 205 has its frequencycoinciding with the frequency of the excitation and its amplitudevarying with an electrostatic capacitance of the capacitor C_(in), thesensor system 205 can be regarded as an amplitude modulation system.

(Operation)

Since input and output properties of the sensor system 205 in FIG. 10and input and output properties of the above-described sensor system 101in FIG. 2 are the same from the above description, operation of thepresent exemplary embodiment is the same as that of Exemplary Embodiment1 or Exemplary Embodiment 2 and therefore description thereof isomitted.

Exemplary Embodiment 3 of the present invention described in theforegoing can be considered to have the following characteristics.Specifically, the electrostatic capacitance sensor 206 of the presentinvention has a first electrode (electrode of the capacitor C_(in) onthe side of the node N1 in FIG. 10), a second electrode (electrode ofthe capacitor C_(in) on the side of the node N2 in FIG. 10), and asensor system 205 configured with a driving circuit (corresponding tothe excitation generating unit 102) which applies a voltage to the firstelectrode and a detecting circuit (I-V converter 207) which measures andoutputs current flowing through the second electrode, in which atouching state or coordinates of an indicator are detected by detectingan electrostatic capacitance of the capacitor C_(in) formed by the firstelectrode and the second electrode.

The electrostatic capacitance sensor 206 of the present inventionincludes the demodulating unit 105 which demodulates an amplitudemodulated signal which is an output of the sensor system 205, and thedemodulating unit 105 generates a demodulated signal using both aresponse of the sensor system 205 in a period when the excitationgenerating unit 102 outputs a sine wave and a response of the sensorsystem 205 in a period, at least either immediately before orimmediately after the aforementioned period, when the excitationgenerating unit 102 does not output a sine wave.

EXAMPLES 1

An electrostatic capacitive touch panel of the present invention will beillustrated.

(Configuration)

FIG. 11 illustrates a configuration of an electrostatic capacitive touchpanel 130 of the present invention. The touch panel illustrated in FIG.11 uses the capacitance of a capacitor C_(in) formed between a fingerand a resistive sheet 131 to detect the presence or absence of a touchand the position of the touch.

An ITO (Indium-Tin-Oxide) film is used for the resistive sheet 131. TheITO film is a solid film having a uniform sheet resistance value, 800ohms in this example, provided on a glass substrate, not depicted. Aninsulator, which is a polarizer 132 used for forming a liquid-crystaldisplay in this example, is attached on the ITO film with an acid-freeadhesive.

Wiring lines are connected to the four corners of the ITO film 131. Thewiring lines are connected to four sensor systems 101 as illustrated inFIG. 11. The configuration of each of the sensor systems is the same asthat in the embodiment 1. Each of the four sensor systems takes an inputof an output voltage from an excitation generating unit 102 and anoutput from each of the sensor systems is provided to an associated oneof demodulating units 105 (demodulating units 0 to 3).

Outputs from the demodulating units are provided to a block, notdepicted, including a signal processing circuit and the presence orabsence of a touch and the position of the touch is calculated in theblock including the signal processing circuit on the basis of theoutputs from the demodulating units.

(Operation)

Operations of the electrostatic capacitive touch panel of the presentinvention will be illustrated with reference to FIG. 12.

The electrostatic capacitive touch panel of the present invention isassembled on the display surface of a liquid-crystal display (LCD) andis driven in such a manner that LCD drive noise is avoided.

A non-addressing indication signal in FIG. 12 is a signal thatexplicitly indicates a non-addressing period of the LCD and is high in anon-addressing period. The term non-addressing period herein refers to aperiod during which the scan lines of the LCD are not scanned and is theperiod from the end of selection of the last scan line to selection ofthe first scan line.

One of the features of the drive of the present invention is that thereis a period (t₁ to t_(1′)) during which a sinusoidal wave is providedfor excitation to sense a touch during a non-addressing period and thereare periods (t₀ to t_(0′) and t₂ to t_(2′)) during which the sinusoidalwave is stopped and noise is acquired.

Since noise is acquired during a non-addressing period, the noiseincludes external noise but does not include LCD drive noise.Consequently, noise present in a period (t₁ to t_(1′)) in which a touchis sensed can be precisely estimated and removed.

The excitation generating unit 102 generates an intermittent sinusoidalvoltage as illustrated as the second waveform from the top of FIG. 12.The sinusoidal voltage is used for excitation of the sensor systems. Inorder to obtain the excitation generating unit output voltage in FIG.12, the excitation generating unit is provided with a sinusoidal wavewith a frequency of 100 kHz and an amplitude of 1.5 V_(pp) (1.5 voltspeak-to-peak) from a sinusoidal wave generating unit 103 and a DCvoltage of DC=1.2 V from an DC generating unit 104. The excitationgenerating unit outputs an intermittent sinusoidal voltage with anoffset of 1.2 V, a frequency of 100 kHz and an amplitude of 1.5 V_(pp).In a period during which the sinusoidal wave is stopped, the excitationgenerating unit outputs a voltage of DC=1.2 V.

The voltage generated by the excitation generating unit is provided tothe four sensor systems 101, which are herein referred to as the sensorsystem of ch0, the sensor system of ch1, the sensor system of ch2, andthe sensor system of ch3 for convenience. The voltage generated by theexcitation generating unit 102 is provided to a non-inverting inputterminal of an operational amplifier 110 in each sensor system and thevoltage appears at an inverting input terminal due to an imaginary shortoperation of the operational amplifier. Specifically, when theexcitation generating unit 102 outputs a voltage with a frequency of 100kHz and an amplitude of 1.5 V_(pp), the voltage with a frequency 100 kHzand an amplitude of 1.5 V_(pp) is applied to the ITO 131.

When the capacitance of a capacitor C_(in) is formed, an AC currentflows to the human body from each sensor system through correspondingconductance G₀ to G₃, which is determined according to the position ofthe finger, and the capacitor C_(in).

An output from each sensor system is the intermittent sinusoidal voltageon which noise is superimposed and the amplitude of the output isdetermined by the magnitude of the AC current. The sensor system of ch1is chosen as a representative example from among the sensor systems andthe output voltage of the sensor system of ch1 is illustrated as f₁(t)in FIG. 12.

An operation of the demodulating unit 105 will be illustrated by takingch1 as an example.

The demodulating unit 105 b of ch1 uses signals y_(n)(t), x_(n)(t) andz_(n)(t), where n is an integer, of the output voltage f₁(t) from thesensor system of ch1, as illustrated in FIG. 12, to output the amplitudeD₁(t) of a true signal of x_(n)(t).

In the demodulating unit 105 b, the output voltage f₁(t) from the sensorsystem is sampled at sampling intervals Δt=0.4 microseconds to obtainf₁(aΔt), where a is an integer sample number.

x₁(iΔt), y₁(iΔt), z₁(iΔt) are signals extracted, respectively, fromf₁(aΔt) as follows: y₁(iΔt), where i=0 to 399, was set as f(aΔt), wherea=3801 to 4200; x₁(iΔt), where i=0 to 1624, was set as f(aΔt), wherea=4251 to 5875; and z₁(iΔt), where i=0 to 399, was set as f(aΔt), wherea=6001 to 6400.

In this example, each of periods corresponding to y₁(t) and z₁(t) isdivided into four segments and a vector of a 100-kHz component iscalculated for each of the segments in order to accurately estimate aphase rotation of noise.

Specifically, the calculations are illustrated by formulas 18 to 25given below.

$\begin{matrix}{Y_{1,1} = {\frac{1}{100}{\sum\limits_{i = 0}^{99}{y\left\{ {(i\;)\Delta\; t} \right\}{\exp\left( {{- j}\; 2\pi\; 100\mspace{11mu}{kHz}\mspace{11mu} i\;\Delta\; t} \right)}}}}} & \left\lbrack {{Formula}\mspace{14mu} 18} \right\rbrack \\{Y_{1,2} = {\frac{1}{100}{\sum\limits_{i = 0}^{99}{y\left\{ {\left( {i\; + 100} \right)\Delta\; t} \right\}{\exp\left( {{- j}\; 2\pi\; 100\mspace{11mu}{kHz}\mspace{11mu} i\;\Delta\; t} \right)}}}}} & \left\lbrack {{Formula}\mspace{14mu} 19} \right\rbrack \\{Y_{1,3} = {\frac{1}{100}{\sum\limits_{i = 0}^{99}{y\left\{ {\left( {i + 200}\; \right)\Delta\; t} \right\}{\exp\left( {{- j}\; 2\pi\; 100\mspace{11mu}{kHz}\mspace{11mu} i\;\Delta\; t} \right)}}}}} & \left\lbrack {{Formula}\mspace{14mu} 20} \right\rbrack \\{Y_{1,4} = {\frac{1}{100}{\sum\limits_{i = 0}^{99}{y\left\{ {\left( {i + 300}\; \right)\Delta\; t} \right\}{\exp\left( {{- j}\; 2\pi\; 100\mspace{11mu}{kHz}\mspace{11mu} i\;\Delta\; t} \right)}}}}} & \left\lbrack {{Formula}\mspace{14mu} 21} \right\rbrack \\{Z_{1,1} = {\frac{1}{100}{\sum\limits_{i = 0}^{99}{z\left\{ {(i\;)\Delta\; t} \right\}{\exp\left( {{- j}\; 2\pi\; 100\mspace{11mu}{kHz}\mspace{11mu} i\;\Delta\; t} \right)}}}}} & \left\lbrack {{Formula}\mspace{14mu} 22} \right\rbrack \\{Z_{1,2} = {\frac{1}{100}{\sum\limits_{i = 0}^{99}{z\left\{ {\left( {i\; + 100} \right)\Delta\; t} \right\}{\exp\left( {{- j}\; 2\pi\; 100\mspace{11mu}{kHz}\mspace{11mu} i\;\Delta\; t} \right)}}}}} & \left\lbrack {{Formula}\mspace{14mu} 23} \right\rbrack \\{Z_{1,3} = {\frac{1}{100}{\sum\limits_{i = 0}^{99}{z\left\{ {\left( {i\; + 200} \right)\Delta\; t} \right\}{\exp\left( {{- j}\; 2\pi\; 100\mspace{11mu}{kHz}\mspace{11mu} i\;\Delta\; t} \right)}}}}} & \left\lbrack {{Formula}\mspace{14mu} 24} \right\rbrack \\{Z_{1,4} = {\frac{1}{100}{\sum\limits_{i = 0}^{99}{z\left\{ {\left( {i\; + 300} \right)\Delta\; t} \right\}{\exp\left( {{- j}\; 2\pi\; 100\mspace{11mu}{kHz}\mspace{11mu} i\;\Delta\; t} \right)}}}}} & \left\lbrack {{Formula}\mspace{14mu} 25} \right\rbrack\end{matrix}$

Then, the amplitudes and phases of forward noise and backward noise areobtained.

The mean value of the amplitudes of the segments is calculated first asfollows. The amplitude |Y_(m)| of forward noise and the amplitude|Z_(m)| of backward noise are each calculated as:|Y _(m)|=(|Y _(1,1) |+|Y _(1,2) |+|Y _(1,3) |+|Y _(1,4)|)/4  [Formula26]|Z _(m)|=(|Z _(1,1) |+|Z _(1,2) |+|Z _(1,3) |+|Z _(1,4)|)/4  [Formula27]The phase of each segment is calculated from the results of thecalculations of Formulas 18 to 25 as follows.

angle [Y_(1, 1)], angle [Y_(1, 2)], angle [Y_(1, 3)], angle [Y_(1, 4)],and angle [Z_(1, 1)] angle [Z_(1, 2)], angle [Z_(1, 3)], angle[Z_(1, 4)]. Here, angle [Y_(1, 1)] represents the phase of Y_(1, 1).

The phases calculated above are limited within the range of ±π. Sincethis is inconvenient for phase estimation, 2nπ, where n is an integer,is added as appropriate to smoothly link the phases.

This operation can be better understood from observation of actualshifts of phase of a 100-kHz component of a sensor system outputcontaining external noise from an inverter circuit of a fluorescentlamp.

FIG. 13 illustrates waveforms obtained by driving an electrostaticcapacitive touch panel of the present invention located near an invertercircuit of a fluorescent lamp. The chart at the top represents thevoltage of the ITO, the second one from the top represents a waveformobtained by sampling a sensor system output of ch1, the third onerepresents the amplitude of a 100-kHz component calculated from eachsegment including 100 samples, and the chart at the bottom representsthe phase of the 100-kHz component calculated from each segmentincluding 100 samples. The chart at the bottom is the result of additionof 2nπ, where n is an integer, to the phases limited within the range of±π to smoothly link the phases.

The result shows smooth phase variations and it can be seen from theresult that the phases can be smoothly linked by adding 2nπ asappropriate, where n is an integer.

Further, the gradients of the four phases angle [Y_(1, 1)], angle[Y_(1, 2)], angle [Y_(1, 3)] and angle [Y_(1, 4)] obtained from theforward noise and angle [Z_(1, 1)], angle [Z_(1, 2)], angle [Z_(1, 3)]and angle [Z_(1, 4)] obtained from the backward noise are used toestimate in which direction the phase has rotated during the period fromthe forward noise to the backward noise, and to what degree.

Let angle [Y_(1, 1)]′, angle [Y_(1, 2)]′, angle [Y_(1, 3)]′, angle[Y_(1, 4)]′ and angle [Z_(1, 1)]′, angle [Z_(1, 2)]′, angle [Z_(1, 3)]′,angle [Z_(1, 4)]′ denote phases that have undergone the two processesillustrated above, i.e. the process for removing the limitations of therange of ±π and the process for estimating the direction and degree ofrotation from the gradients of the phases of the forward noise and thebackward noise. Then the phase angle [Y_(m)] of the forward noise andthe phase angle [Z_(m)] of the backward noise are calculated as follows.angle[Y _(m)]=(angle[Y _(1,1)]′+angle[Y _(1,2)]′+angle[Y_(1,3)]′+angle[Y _(1,4)]′)/4  [Formula 28]angle[Z _(m)]=(angle[Z _(1,1)]′+angle[Z _(1,2)]′+angle[Z_(1,3)]′+angle[Z _(1,4)]′)/4  [Formula 29]

Note that it can also be seen from the third chart from the top of FIG.13 that the amplitude of noise present in the period x(t) can beestimated by approximation by linking the forward noise and the backwardnoise by straight lines.

Vector Y_(m) is determined by |Y_(m)| and angle [Y_(m)] obtained aboveand vector Z_(m) is determined by |Z_(m)| and angle [Z_(m)] obtainedabove.

Then, noise vectors Y and Z at time instants t₁ and t_(1′) are estimatedfrom Y_(m) and Z_(m) according to the procedure described in theembodiment 2.

Then, the mean vector M₁ of the vectors Y and Z is calculated from thevectors Y and Z according the procedure described in the embodiment 2.

Further, vector X₁ is obtained and X₁−M₁ is calculated. X can be writtenas the following formula.

$\begin{matrix}{X_{1} = {\frac{1}{1625}{\sum\limits_{i = 0}^{1624}{x\left\{ {i\;\Delta\; t} \right\}{\exp\left( {{- j}\; 2\;\pi\; 100\;{kHz}\mspace{11mu} i\;\Delta\; t} \right)}}}}} & \left\lbrack {{Formula}\mspace{14mu} 30} \right\rbrack\end{matrix}$

Here, Δt is the sampling interval and j is an imaginary unit.

The magnitude |X₁−M₁| of the vector X₁−M₁ is output as the output D₁(t)from the demodulating unit 105 b as illustrated in FIG. 12

In the next non-addressing period, |X₂−M₂| is calculated similarly fromy₂(t), x₂(t) and z₂(t) and is output from the demodulating unit asillustrated in FIG. 12.

In this way, |X_(n)−M_(n)| is calculated from y_(n)(t), x_(n)(t) andz_(n)(t) and is output from the demodulating unit.

A block diagram of a signal processing unit for obtaining Y_(1, 1),Y_(1, 2), Y_(1, 3), . . . , X₁, . . . , Z_(1, 3), Z_(1, 4) from anoutput voltage f₁(t) of the sensor system illustrated above will beillustrated with reference to FIG. 14.

Output f(t) of the sensor system 101 in FIG. 14 corresponds to theoutput voltage f₁(t) of the sensor system illustrated above withreference to FIG. 11. f(t) is provided to a sampler 140, in which f(t)is converted to a discrete-time signal f(aΔt), where a=0, 1, 2, . . . ,with time intervals Δt=0.4 microseconds. Then f(aΔt) is input into twomultipliers (multiplier I 141 a and multiplier Q 141 b). The multiplierI 141 a sequentially multiplies f(aΔt) by cos (ωaΔt), where a=0, 1, 2,3, . . . and ω=2π100 kHz, and sequentially outputs the result at timeintervals Δt. Similarly, the multiplier Q 141 b sequentially multipliesf(aΔt) by sin (ωaΔt), where a=0, 1, 2, 3, . . . and ω=2π100 kHz andsequentially outputs the result at time intervals Δt.

An output of a sinusoidal wave generating unit 103 is used as cos (ωaΔt)in the multiplier I; a signal obtained by converting the output from thesinusoidal generating unit by passing through a −90-degree phase-shifter145 is used as sin (ωaΔt) in the multiplier Q.

The outputs from the multipliers I 141 a and Q 141 b are input intointegrators I 142 a and Q 142 b, respectively, and the integrators add asignal input in a period during which a control signal provided from acontroller 146 is active.

For example, to obtain Y1, 1, the controller provides an active signalto the integrators in a period during which the value of a in f(aΔt) is3801 to 3900. This causes the integrator I 142 a to calculate

$\begin{matrix}{{\sum\limits_{a = 3801}^{3900}{{f\left( {a\;\Delta\; t} \right)}{\cos\left( {\omega\; a\;\Delta\; t} \right)}}} = {\sum\limits_{i = 0}^{99}{{y_{1}\left( {i\;\Delta\; t} \right)}{\cos\left( {\omega\; a\;\Delta\; t} \right)}}}} & \left\lbrack {{Formula}\mspace{14mu} 31} \right\rbrack\end{matrix}$

In other words, a value which is one hundred times that of the real partof Y_(1, 1) in formula 17 is calculated.

Signals integrated in a predetermined time period taken into a registerI 143 a and a register Q 14 n 3 b and are multiplied by 1/N (N is thenumber of integrated samples) by multipliers 144 connected to theregisters.

Through this process, the multiplier I 144 a sequentially outputs thereal parts of Y_(1, 1), Y_(1, 2), Y_(1, 3), . . . , X₁, . . . ,Z_(1, 3), Z_(1, 4), i.e. the values of Re{Y_(1, 1)}, Re{Y_(1, 2)},Re{Y_(1, 3)}, . . . , Re{X₁}, . . . , Re{Z_(1, 3)}, Re{Z_(1, 4)}, andthe multiplier Q 144 b sequentially outputs the imaginary parts ofY_(1, 1), Y_(1, 2), Y_(1, 3), . . . , X₁, . . . , Z_(1, 3), Z_(1, 4)multiplied by −1, i.e. the values of −Im{Y_(1, 1)}, −Im{Y_(1, 2)},−Im{Y_(1, 3)}, . . . , −Im{X₁}, . . . −Im{Z_(1, 3)}, −Im{Z_(1, 4)}.

These values are sequentially input into a computer, not depicted, inwhich the magnitudes and phases are calculated.

Results of an experiment on noise removal using the present inventionand conventional noise removal, i.e. noise removal using only frequencyseparation will now be described.

For the experiment, a touch panel in FIG. 11 was provided and aninverter circuit of an inverter fluorescent lamp was placed 30 cm abovethe touch panel. Outputs from sensor systems were observed and it wasfound that noise from the inverter circuit is obviously present in theoutputs.

The measurement was made for approximately 10 seconds and, approximately5 seconds after the start of the measurement, the center of the touchpanel was touched with a finger. Results of the experiment areillustrated in FIG. 15.

FIG. 15 (b) illustrates the result of the experiment with the presentinvention. Each of |X_(n)−M_(n|), which was an output of D₁(t), wasplotted as one point and 653 points were linked by straight lines.

On the other hand, FIG. 15 (a) illustrates the result of the experimenton noise removal using only frequency separation. Specifically, theamplitude of a 100-kHz component of an output signal from the sensorsystem in a period of a sinusoidal wave with a 100 kHz excitation wasobtained by |X_(n)−0|.

It has been confirmed that implementation of the present inventionachieves a 9 dB improvement in signal-to-noise ratio, from conventional1.36 to 3.87 in the present invention, where the signal S is themagnitude of signal difference between the presence and absence of atouch and the noise N is the standard deviation in the absence of atouch.

EXAMPLE 2

Example 2 is a new technique applicable to Example 1. The presentExample of the present invention discloses a technique for obtaining ahigh S/N ratio with respect to various noises.

In Example 1, with respect to every n (n denotes a number 1, 2, 3, . . .sequentially applied to a non-address period), |X_(n)−M_(n)| wascalculated and an obtained value was taken as an output of thedemodulating unit.

In Example 1, however, when calculation of |X_(n)−M_(n)| is applied toevery n, an S/N ratio might be reduced in some cases. For example, inExample 1, this is the case of a signal obtained with operation of theinverter circuit stopped. Stopping the inverter circuit removes explicitnoise source to make other external noise conspicuous as noise. In theexperiment in which the inverter circuit was stopped, when the output ofthe demodulating unit was simply set to |X_(n)|, the S/N ratio was 52.3and when the output of the demodulating unit was set to |X_(n)−M_(n)|,the S/N ratio was reduced to 30.2.

As a result of analyses by the inventors, it was found that when theabove-described other noise was major noise, a change in the phasesobtained in Example 1, i.e. angle [Y_(n,1)]′, angle [Y_(n,2)]′, angle[Y_(n,3)]′ and angle [Y_(n,4)]′, and angle [Z_(n,1)]′, angle [Z_(n,2)]′,angle [Z_(n,3)]′ and angle [Z_(n,4)]′ (the first subscript n denotes anumber 1, 2, 3, . . . sequentially applied to a non-address period andthe second subscript denotes a segment number) was so large that M_(n)calculated using them did not properly reflect noise included in X_(n).

Noting a phase change, a control means which sets a vector M_(n) to zerowhen the phase change is larger than a predetermined value is thus addedto the demodulating unit. A flow chart of processing in the demodulatingunit is illustrated in FIG. 16. Operation of the demodulating unit willbe described along the flow chart of FIG. 16.

(Step 1)

According to the method described in Example 1, calculation is executedof phases of four segments of the forward noise, angle [Y_(n,1)], angle[Y_(n,2)], angle [Y_(n,3)] and angle [Y_(n,4)], and phases of foursegments of the backward noise, angle [Z_(n,1)], angle [Z_(n,2)], angle[Z_(n,3)] and angle [Z_(n,4)]. The first subscript n denotes a number 1,2, 3, . . . sequentially applied to a non-address period and the secondsubscript denotes a segment number.

A specific example will be described. Among the values obtained by theexperiment, illustrated in FIG. 17 and FIG. 18 is an example where thenumber n of the non-address period is 1. FIG. 17 illustrates phases,angle [Y_(1,s)] and angle [Z_(1,s)], obtained when the inverter circuitis operated, with s representing a segment number, and FIG. 18illustrates phases obtained when the inverter circuit is stopped.

(Step 2)

Since the range of phases obtained at STEP 1 is from −π to π, processingfor removing the limitation is executed similarly to Example 1. Inaddition, from inclinations of the phases of the forward noise and thebackward noise, rotation direction and amount of the phases areestimated. The phases subjected to the processing are assumed to beangle [Y_(n,1)]′, angle [Y_(n,2)]′, angle [Y_(n,3)]′ and angle[Y_(n,4)]′, and angle [Z_(n,1)]′, angle [Z_(n,2)]′, angle [Z_(n,3)]′ andangle [Z_(n,4)]′.

A specific example will be described. Results of the processing of STEP2 executed with respect to the phases obtained at STEP 1 are illustratedin FIG. 19 and FIG. 20. FIG. 19 illustrates phases, angle [Y_(1, s)]′and angle [Z_(1,s)]′, obtained when the inverter circuit is operated,with s representing a segment number, and FIG. 20 illustrates a phaseobtained when the inverter is stopped.

(Step 3)

With respect to the above-obtained phases, six phase differences definedin the following are obtained.dY(s)=angle[Y _(n,s+1)]′−angle[Y _(n,s) ]′,s=1,2,3  [Formula 32]dZ(s)=angle[Z _(n,s+1)]′−angle[Z _(n,s) ]′,s=1,2,3  [Formula 33]

Next, absolute values of these six phase differences are obtained toobtain their maximum value.

Indicated in Table 1 are results of maximum absolute values of the phasedifferences obtained with respect to the phase data in FIG. 19 when theinverter circuit is on and with respect to the phase data in FIG. 20when the inverter circuit is off.

TABLE 1 inverter circuit ON inverter circuit OFF 0.40 (rad) 2.11 (rad)(Step 4)

Using “the maximum absolute values of the phase differences” obtained atSTEP 3, the output value of the demodulating unit is controlledaccording to the following description.If (the maximum absolute value of the phase difference)>the controlthreshold, then output=|X _(n)−0|.  [Formula 34]If (the maximum absolute value of the phase difference)≤the controlthreshold, then output=|X _(n) −M _(n)|.  [Formula 35]

In the present Example, a specific value as a control threshold was setto be 1.0 (rad). Accordingly, the output of the demodulating unit withrespect to data in the non-address period whose number n is =1 isexpressed as follows.Output value of the demodulating unit=|X ₁ −M ₁| when the inverter ison.  [Formula 36]Output value of the demodulating unit=|X ₁−0|=|X ₁| when the inverter isoff.  [Formula 37]

The above control at STEP 4 can be described as follows when using acoefficient k.If (the maximum absolute value of the phase difference)>the controlthreshold, then k=0.  [Formula 38]If (the maximum absolute value of the phase difference)≤the controlthreshold, then k=1.  [Formula 39]Output=|X _(n) −k·M _(n)|.  [Formula 40]

The above-described STEP 1 to STEP 4 is applied to each number n of thenon-address period. The value of the above coefficient k is determinedby a large/small relation between “a control threshold” and “a maximumabsolute value of a phase difference” as described above. Then, thephase difference is obtained from the forward noise and the backwardnoise. The forward noise and the backward noise here are outputs of thesensor system in the period when the excitation generating unit stopswave transmission. In other words, the value of k is determined using aresponse of the sensor system in the period when the excitationgenerating unit does not output a sine wave.

The effects of the present Example were evidenced by the followingexperimental results. Similarly to Example 1, the experiment has beenexecuted by the method of driving the touch panel in FIG. 11 to evaluatean S/N ratio of an output value of the demodulating unit. How the S/Nratio changes with a value to be applied to the above control thresholdwas obtained by the experiment. FIG. 21 illustrates experimental resultsobtained in an environment in which the inverter circuit was operated.With reference to FIG. 21, it can be found that when the controlthreshold is set to 0.7 (rad) or above, a high S/N ratio (3.87) isobtained. When the control threshold is set to 0, the S/N ratio isreduced to 1.36. FIG. 22 illustrates experimental results obtained in anenvironment in which the inverter circuit was stopped. With reference toFIG. 22, it can be found that when the control threshold is set to 1.2(rad) or below, a high S/N ratio (52.3) is obtained. Accordingly, thehighest S/N ratio can be obtained both in the environment in which theinverter circuit is operated and the environment in which the invertercircuit is stopped by setting the control threshold to be 0.7 or aboveand 1.2 or below. In other words, applying the control means of Example2 to Example 1 avoided the problem of reduction in an S/N ratio in theenvironment in which the inverter circuit was stopped, while increasingthe S/N ratio in the environment in which the inverter circuit wasoperated.

EXAMPLE 3

Example 3 is other new technique applicable to Example 1. In Example 1,with respect to every n (n denotes a number 1, 2, 3, . . . sequentiallyapplied to a non-address period), |X_(n)−M_(n)| was calculated and anobtained value was taken as an output of the demodulating unit. On theother hand, in Example 2, a coefficient k was calculated with respect toeach n and |X_(n)−k·M_(n)| was taken as an output of the demodulatingunit. The inventors found another method for determining the coefficientk.

In other words, in the present Example, an amplitude of the forwardnoise (noise obtained at the time of non-output of a sine wave beforethe excitation generating unit outputs a sine wave) and an amplitude ofthe backward noise (noise obtained at the time of non-output of a sinewave after the excitation generating unit outputs a sine wave) are usedin order to determine a value of the coefficient k. Specifically, notinga mean value of the forward noise amplitude and the backward noiseamplitude, the demodulating unit is provided with a means for setting avector M_(n) to zero when the mean value is larger than a predeterminedvalue. FIG. 23 illustrates a flow chart of the processing in thedemodulating unit. Operation of the demodulating unit will be describedalong the flow chart in FIG. 23.

(Step 1)

A mean value of the forward noise amplitude |Y_(m)| and the backwardnoise amplitude |Z_(m)| as described in Example 1 is calculated as(|Y_(m)|+|Z_(m)|)/2.

(Step 2)

Using “the mean value of the forward noise amplitude and the backwardnoise amplitude” obtained at STEP 1, the output value of thedemodulating unit is controlled according to the following description.If (the mean value of the forward noise amplitude and the backward noiseamplitude)<the second control threshold, then output=|X_(n)−0|.  [Formula 41]If (the mean value of the forward noise amplitude and the backward noiseamplitude)≥the second control threshold, then output=|X _(n) −M_(n)|,  [Formula 42]the output of the demodulating unit can be expressed as follows with thecoefficient k.If (the mean value of the forward noise amplitude and the backward noiseamplitude)<the second control threshold, then k=0.If (the mean value of the forward noise amplitude and the backward noiseamplitude)≥second control threshold, then k=1.Output=|X _(n) −k·M _(n)|.  [Formula 43]

The above-described STEP 1 to STEP 2 is applied to each number n of thenon-address period. The above coefficient k is determined by alarge/small relation between “a second control threshold” and “a meanvalue of the forward noise amplitude and the backward noise amplitude”as described above. Then, the mean value is obtained from the forwardnoise and the backward noise. The forward noise and the backward noisehere are outputs of the sensor system in the period when the excitationgenerating unit stops wave transmission. In other words, the value of kis determined using a response of the sensor system in the period whenthe excitation generating unit does not output a sine wave.

In addition, it is self-evident that the Output of [Formula 43] canexpress Output=C·|X_(n)−k·M_(n)| (C is a constant) generally.

The effects of the present Example were evidenced by the followingexperimental results. Similarly to Example 1, the experiment has beenexecuted by driving the touch panel in FIG. 11 to evaluate an S/N ratioof an output value of the demodulating unit. How the S/N ratio changeswith a value to be applied to the above second control threshold wasobtained by the experiment. FIG. 24 illustrates experimental resultsobtained in an environment in which the inverter circuit was operated.With reference to FIG. 24, it can be found that when the second controlthreshold is set to 0.02 (V) or below, a high S/N ratio (3.87) isobtained. When the second control threshold is set to 0.06, the S/Nratio is reduced to 1.36. FIG. 25 illustrates experimental resultsobtained in an environment in which the inverter circuit was stopped.With reference to FIG. 25, it can be found that when the second controlthreshold is set to 0.005 (V) or above, a high S/N ratio (52.3) isobtained. Accordingly, the highest S/N ratio can be obtained both in theenvironment in which the inverter circuit is operated and theenvironment in which the inverter circuit is stopped by setting thesecond control threshold to be 0.005 or above and 0.02 or below. Inother words, applying the control means of Example 3 to Example 1avoided the problem of reduction in an S/N ratio in the environment inwhich the inverter circuit was stopped, while increasing the S/N ratioin the environment in which the inverter circuit was operated.

The inventor tried several touch panels configured based on therecitation of Example 2 and Example 3 to evaluate S/N ratios. As aresult, it was experienced that in a case of the experimental productbased on Example 2, none of common control thresholds might exist withwhich the highest S/N ratios could be obtained both in the environmentin which the inverter circuit was operated and the environment in whichthe inverter circuit was stopped in some cases. In such a case,replacing the demodulating unit by that recited in Example 3 resulted inhaving a common second control threshold with which the highest S/Nratios could be obtained both in the environment in which the invertercircuit was operated and the environment in which the inverter circuitwas stopped. In other words, it can be considered that while Example 2and Example 3 have the common effect of obtaining a high S/N ratio withrespect to various noises, Example 3 might be more excellent inobtaining a high S/N ratio in some cases.

EXAMPLE 4

Example 2 recites an example where the coefficient k is determined basedon “a maximum absolute value of a phase difference” and Example 3recites an example where the coefficient k is determined based on “amean amplitude of a forward noise amplitude and a backward noiseamplitude”. Thus, two physical quantities are described for determiningthe coefficient k. Example 4 recites one example for determining thecoefficient k using these two physical quantities. FIG. 26 illustrates aflow chart of processing executed in the demodulating unit. Operation ofthe demodulating unit will be described along the flow chart in FIG. 26.

(Step 1)

This step is the same as STEP 1 of Example 3, which is to obtain a meanvalue of a forward noise amplitude and a backward noise amplitude.

(Step 2)

Similarly to STEP 2 of Example 3, “a mean value of the forward noiseamplitude and the backward noise amplitude” and the second controlthreshold are compared to find a large/small relation therebetween, andwhen the mean value is smaller than the second control threshold,|X_(n)| is output to the demodulating unit to end the processing andwhen the mean value is not less than the second control threshold, theprocessing proceeds to STEP 3.

(Step 3 to Step 6)

These steps are the same as STEP 1 to STEP 4 of Example 2, in which whena maximum absolute value of a phase difference is larger than a controlthreshold, |X_(n)| is output as the output of the demodulating unit toend the processing and when the maximum absolute value of the phasedifference is not more than the control threshold, |X_(n)−M_(n)| isoutput to the demodulating unit to end the processing.

The foregoing STEP 1 to STEP 6 is applied to each number n of thenon-address period.

Expressing an output of the demodulating unit using the coefficient k as|X_(n)−k·M_(n)| leads to determination of the value of k in the presentExample in the following manner. In other words, “a mean value of theforward noise amplitude and the backward noise amplitude” and a secondcontrol threshold are compared to find a large/small relationtherebetween, and when the means value is smaller than the secondcontrol threshold, k is set to zero. In addition, “the mean value of theforward noise amplitude and the backward noise amplitude” and the secondcontrol threshold are compared to find a large/small relationtherebetween, and when the mean value is not less than the secondcontrol threshold and a maximum absolute value of a phase difference islarger than the control threshold, k is set to zero. “The mean value ofthe forward noise amplitude and the backward noise amplitude” and thesecond control threshold are compared to find a large/small relationtherebetween, and when the mean value is not less than the secondcontrol threshold and the maximum absolute value of a phase differenceis not more than the control threshold, k is set to 1.

EXAMPLE 5

Other electrostatic capacitive touch panel of the present invention willbe described.

(Structure)

FIG. 27 illustrates an electrostatic capacitive touch panel of thepresent invention. The present touch panel has a transparent substrate202 on which a plurality of electrodes (H1: 203 a, H2: 203 b, H3: 203 cand V1: 203 d, V2: 203 e, V3: 203 f) are disposed. The present touchpanel detects touching or non-touching and a touch position by detectingan own electrostatic capacitance of an individual electrode, i.e.electrostatic capacitance formed with the ground by the individualelectrode.

The respective electrodes 203 a to 203 f are connected to nods 1 to 6 ofan analog multiplexer 201 by wiring and the analog multiplexer 201 isconnected to the sensor system 101. A configuration of the sensor systemis the same as that of Exemplary Embodiment 1. To the sensor system 101,an output voltage of the excitation generating unit 102 is input and anoutput of the sensor system 101 is applied to the demodulating unit 105.

The output of the demodulating unit 105 is transmitted to a blockincluding a signal processing circuit not shown, so that in the blockincluding the signal processing circuit, touching or non-touching and atouch position are calculated based on the output value of thedemodulating unit 105.

The analog multiplexer 201, the excitation generating unit 102 and thedemodulating unit 105 are connected to a controller 200 so as to havetheir operation and timing controlled.

The plurality of electrodes (H1, H2, H3 and V1, V2, V3) can beimplemented in other configuration than the above-describedconfiguration, i.e. implemented as an oblong planar electrode disposedon the substrate. They may be microwire electrodes embedded in thesubstrate.

While the number of electrodes is set to be 6 in the present Example,the number can be arbitrarily set, and increasing the number enables thesize of the touch panel to be increased or detection precision to beimproved.

(Operation)

With reference to FIG. 27, operation of the electrostatic capacitivetouch panel of the present invention will be described. An analogmultiplexer selection signal in FIG. 28 indicates a node number selectedby the analog multiplexer 201 in FIG. 27.

One characteristic of drive of the present invention is that a periodwhen one electrode is selected by the analog multiplexer 201 has aperiod (t₁ to t₁′) when a sine wave is applied to the electrode todetect touching and periods (t₀ to t₀′ and t₂ to t₂′) when noise isobtained while stopping a sine wave.

Since only external noise appears in a response in the period when theexcitation generating unit 102 does not output a sine wave, i.e. in asensor system output voltage f(t) in the period when the excitationgenerating unit does not output a sine wave, obtaining this enablesmeasurement of external noise with high precision. Then, making the mostof this characteristic enables highly precise estimation and removal ofnoise to be mixed during the period (t₁ to t₁′) for detecting touching.

The excitation generating unit 102 generates an intermittent sine wavevoltage as illustrated in the second waveform from the top in FIG. 28.The voltage is used as an excitation of the sensor system. In order toobtain an output voltage of the excitation generating unit in FIG. 28,the excitation generating unit is supplied with a sine wave having afrequency of 100 kHz and an amplitude of 1.5 Vpp (1.5 volt peak to peak)by the sine wave generating unit 103 and with a DC voltage of 1.2 V bythe DC generating unit 104. Then, the excitation generating unit 102outputs an intermittent sine wave voltage having an offset of 1.2 V, afrequency of 100 kHz and an amplitude of 1.5 Vpp. In a period when thesine wave is stopped, a DC voltage of 1.2 V is output.

The voltage generated by the excitation generating unit 102 is appliedto the sensor system 101. The voltage generated by the excitationgenerating unit 102 is applied to the non-inverting input terminal ofthe operational amplifier 110 in the sensor system, so that the voltageappears at the inverting input terminal due to imaginary shortingoperation of the operational amplifier. In other words, when theexcitation generating unit 102 outputs a voltage having a frequency of100 kHz and an amplitude of 1.5 Vpp, a voltage having a frequency of 100kHz and an amplitude of 1.5 Vpp is applied to an electrode selected bythe multiplexer 201.

When the electrostatic capacitance C_(in) is formed between a finger andthe electrode as a finger approaches the electrode, alternating currentflows from the sensor system 101 toward the finger through the C_(in).An output of the sensor system 101 is a result obtained by superposingnoise on an intermittent sine wave voltage whose amplitude is determinedaccording to an amount of the alternating current. The output voltage ofsensor system is denoted as f(t) in FIG. 28.

Operation of the demodulating unit 105 will be described. Of the outputvoltage f(t) of the sensor system 101, the demodulating unit 105 outputsan amplitude estimation value D(t) of a true signal of x_(n)(t) by usingsignals y_(n)(t), x_(n)(t) and z_(n)(t), with n being an integer andcorresponding to a node number selected by the analog multiplexer 201 asillustrated in FIG. 28. The amplitude estimation value of the truesignal is denoted as |X_(n)−M_(n)| as detailed in Example 1.

As illustrated in FIG. 28, the controller 200 applies 1, 2, 3, . . . 6to the analog multiplexer selection signal, operation timing of theexcitation generating unit 102 and operation timing of the demodulatingunit 105. Sequential selection of the electrodes by the analogmultiplexer results in sequentially obtaining demodulated signalscorresponding to all the six electrodes, respectively.

Then, in the block including the signal processing circuit not shown,touching or non-touching and a touch position are calculated based onthe output value of the demodulating unit 105. In the example asillustrated in FIG. 28, since |X₂−M₂| has a peak, a possibility isdetected that a touch exists on the electrode H2. Then, when anotherpeak is detected in |X₅−M₅|, for example, possibilities are detectedthat a touch exists on the electrode V2 and that a touch exists at thecenter of the touch panel screen.

For the detection of a touch, such an algorithm is used, in addition tothe above peak detection, as causes no erroneous determination bycalculating a difference from a value obtained when no touch is made orusing characteristics of a temporal change in a signal. Thus, touchingor non-touching and a touch position (coordinates) are detected.

In the foregoing, |X_(n)−M_(n)| described in Example 1 is applied as anoutput of the demodulating unit 105, |X_(n)−k·M_(n)| described inExample 2 to Example 4 may be applied as an output of the demodulatingunit 105.

EXAMPLE 6

Other electrostatic capacitive touch panel of the present invention willbe described. Example 6 illustrates an example of application to amutual projected-capacitive type touch panel.

(Structure)

FIG. 29 illustrates a configuration of an electrostatic capacitive touchpanel of the present invention. In the present touch panel, a pluralityof electrodes (H1: 203 a, H2: 203 b, H3: 203 c and V1: 203 d, V2: 203 e,V3: 203 f) are disposed on the transparent substrate 202. Consequently,nine intersections are formed on the substrate at which the electrodesintersect with each other. At each intersection, electrostaticcapacitance is formed due to two electrodes intersecting with eachother. When an indicator approaches the vicinity of an intersection,nearby lines of electric force change, so that a value of theelectrostatic capacitance formed by the two intersecting electrodeschanges. The present touch panel detects touching or non-touching and atouch position by detecting electrostatic capacitance formed by twointersecting electrodes.

The respective electrodes (203 d, 203 e, 203 f) extending in a lateraldirection are connected respectively to nodes 1 to node 3 of an analogmultiplexer 209 by wiring, and the analog multiplexer 209 is connectedto an output terminal of the excitation generating unit 102. Theelectrodes (203 a, 203 b, 203 c) extending in a vertical direction areone-to-one connected to three I-V converters 207 a, 207 b and 207 c bywiring. A configuration and operation of each of the I-V converters 207a, 207 b and 207 c are the same as those of 207 in Exemplary Embodiment3. An output of the I-V converter is applied to the demodulating unit(105 a, 105 b, 105 c).

The outputs of the demodulating units 105 a to 105 c are transmitted toa block including a signal processing circuit not shown, so that in theblock including the signal processing circuit, touching or non-touchingand a touch position are calculated based on the output values of thedemodulations unit 105 a to 105 c.

The analog multiplexer 209, the excitation generating unit 102 and thedemodulating units 105 a to 105 c are connected to a controller notshown so as to their operation and timing controlled.

In the present Example, although the number of electrodes is set to 6,it can be arbitrarily set and increasing the number enables the size ofthe touch panel to be increased or detection precision to be improved.

(Operation)

With reference to FIG. 30, operation of the electrostatic capacitivetouch panel of the present invention will be described. An analogmultiplexer selection signal in FIG. 30 indicates a node number selectedby the analog multiplexer 209 in FIG. 29.

One characteristic of drive of the present invention is that a periodwhen one electrode is selected among V1, V2 and V3 by the analogmultiplexer 209 has a period (t₁ to t₁′) when a sine wave is applied toa selected electrode to detect touching and periods (t₀ to t₀′ and t₂ tot₂′) when noise is obtained while stopping a sine wave.

Since only external noise appears in an I-V converter output voltage inthe period when the excitation generating unit 102 does not output asine wave, i.e. in a sensor system output voltage in the period when theexcitation generating unit 102 does not output a sine wave, obtainingthis enables measurement of external noise with high precision. Then,making the most of the characteristic enables highly precise estimationand removal of noise to be mixed during the period (t₁ to t₁′) fordetecting touching.

Since in the present Example, the three I-V converters 207 a to 207 cexist, output voltages of these I-V converters are referred to as f₀(t),f₁(t), and f₂(t) to be distinguished. It is assumed here that f₀(t)represents an output voltage of the I-V converter 207 a connected to ademodulating unit 0:105 a, f ₁(t) represents an output voltage of theI-V converter 207 b connected to a demodulating unit 1:105 b and f ₂(t)represents an output voltage of the I-V converter connected to ademodulating unit 2:105 c.

FIG. 30 exemplifies f₁(t) as a representative of the above three. Sincethe output voltage of the I-V converter can be considered as an outputof the sensor system as described in Exemplary Embodiment 3, it isreferred to as a sensor system output voltage in the figure.

The excitation generating unit 102 generates an intermittent sine wavevoltage as illustrated in the second waveform from the top in FIG. 30.The voltage is used as an excitation of the sensor system. In order toobtain an output voltage of the excitation generating unit in FIG. 30,the excitation generating unit 102 is supplied with a sine wave having afrequency of 100 kHz and an amplitude of 1.5 Vpp (1.5 volt peak to peak)by the sine wave generating unit 103 and with a DC voltage of 1.2 V bythe DC generating unit 104. Then, the excitation generating unit 102outputs an intermittent sine wave voltage having an offset of 1.2 V, afrequency of 100 kHz and an amplitude of 1.5 Vpp. In a period when thesine wave is stopped, a DC voltage of 1.2 V is output.

The voltage generated by the excitation generating unit 102 issequentially applied to the plurality of electrodes (V1 to V3)configuring the sensor system through the analog multiplexer 209.

When the electrostatic capacitances C_(in(H1,V1)), C_(in(H1,V2)) . . .C_(in(H3,V3)), with a subscript in a parenthesis assumed to be names ofintersecting electrodes, are formed at nine intersection points at whichthe electrodes intersect with each other, and when a finger approaches aspecific electrode, the value of the electrostatic capacitance isreduced. Then, the amplitude of the I-V converter is responsivelyreduced. The output voltage f₁(t) of the I-V converter 207 b connectedto the demodulating unit 1:105 b is illustrated in FIG. 30.

Operation of the demodulating unit 1 (105 b) will be described. Thedemodulating unit 105 b outputs an amplitude estimation value D₁(t) of atrue signal of x_(n)(t) by using signals y_(n)(t), x_(n)(t) and z_(n)(t)of the output voltage f₁(t) of the sensor system, with n being aninteger and corresponding to a node number selected by the analogmultiplexer 209 as illustrated in FIG. 30. The amplitude estimationvalue of the true signal is denoted as |X_(n)−M_(n)| as detailed inExample 1.

As illustrated in FIG. 30, the controller not shown applies, to theanalog multiplexer selection signal, 1,2,3, 1,2,3, . . . , and alsooperation timing of the excitation generating unit 102 and operationtiming of the demodulating unit 105 a to 105 c. Sequential selection ofthe electrodes by the analog multiplexer 209 results in sequentiallyapplying an excitation to all the three electrodes V1, V2 and V3. On theother hand, pairs each composed of the I-V converter and thedemodulating unit are connected to the three electrodes H1, H2 and H3,respectively, to output demodulated signals in parallel. Therefore, whenthe analog multiplexer finishes sequential selection, 1, 2, 3,demodulated signals corresponding to the electrostatic capacitances atall the nine intersection points can be obtained.

Then, in the block including the signal processing circuit not shown,touching or non-touching and a touch position are calculated based onthe output values of the demodulating units 105 a to 105 c. In theexample as illustrated in FIG. 30, a peak (minimum) is seen in ademodulated signal obtained when the analog multiplexer selects the node2, i.e. |X₂−M₂|. This enables detection of a possibility that a touch ismade at the intersection point between the electrode H2 and theelectrode V2.

For the detection of a touch, such an algorithm is used, in addition tothe above peak detection, as causes no erroneous determination bycalculating a difference from a value obtained when no touch is made orusing characteristics of a temporal change in a signal. Thus, touchingor non-touching and a touch position (coordinates) are detected.

In the foregoing, |X_(n)−M_(n)| described in Example 1 is applied as anoutput of the demodulating units 105 a to 105 c, |X_(n)−k·M_(n)|described in Example 2 to Example 4 may be applied as an output of thedemodulating units 105 a to 105 c.

It can be considered that the above-described Example 4 of the presentinvention has the following characteristics. Specifically, Example 4includes a first electrode (i.e. the electrode V2), a second electrode(i.e. the electrode H2), and a sensor system configured with a drivingcircuit (corresponding to the excitation generating unit 102) whichapplies a voltage to the first electrode and a detecting circuit (207)which measures and outputs current flowing through the second electrode,and detects electrostatic capacitance of a capacitor formed by the firstelectrode and second electrode, thereby detecting a touching state orcoordinates of an indicator.

In addition, the touch panel of the present invention includes thedemodulating units 105 a to 105 c which demodulate an amplitudemodulated signal as an output of the sensor system 205, in which thedemodulating units 105 a to 105 c generate a demodulated signal usingboth a response of the sensor system in a period when the excitationgenerating unit 102 outputs a sine wave and a response of the sensorsystem in a period, at least either immediately before or immediatelyafter the aforementioned period, when the excitation generating unitdoes not output a sine wave.

INDUSTRIAL APPLICABILITY

The present invention is applicable to electronic devices that use anamplitude modulation and demodulation system, such as electrostaticcapacitance sensors, touch panels, touch sensors, and the like.

DESCRIPTION OF SYMBOLS

-   -   100: Electrostatic capacitance sensor    -   101: Sensor system    -   102: Excitation generating unit    -   103: Sinusoidal wave generating unit    -   104: DC generating unit    -   105, 105 a, 105 b, 105 c, 105 d: Demodulating unit    -   110: Operational amplifier    -   111: Adder    -   120: Electronic device    -   130: Electrostatic capacitive touch panel    -   131: Resistive sheet (ITO)    -   132: Polarizer    -   140: Sampler    -   141: Multiplier, 141 a: Multiplier I, 141 b: Multiplier Q    -   142: Integrator, 142 a: Integrator I, 142 b: Integrator Q    -   143: Register, 143 a: Register I, 143 b: Register Q    -   144: Multiplier, 144 a: Multiplier I, 144 b: Multiplier Q    -   145: Phase shifter    -   146: Controller    -   200: controller    -   201: multiplexer    -   202: substrate    -   203 a, 203 b, 203 c, 203 d, 203 e, 203 f: electrode    -   205: sensor system    -   206: electrostatic capacitance sensor    -   207: current voltage converter (I-V converter)    -   208: DC bias circuit    -   209: multiplexer

The invention claimed is:
 1. An electronic apparatus comprising: asensor system; an excitation generating unit which generates anintermittent alternating current (AC) wave or square wave and appliesthe intermittent alternating current wave or square wave to the sensorsystem; and a demodulating unit which demodulates an amplitude modulatedsignal which is an output of the sensor system, wherein the demodulatingunit generates a demodulated signal using both a response of the sensorsystem in a period when the excitation generating unit outputs analternating current wave or square wave and a response of the sensorsystem in a period, at least either immediately before or immediatelyafter the aforementioned period, when the excitation generating unitdoes not output an alternating current wave or square wave, and whereinwith a vector as X, which is obtained from an amplitude and a phase, theamplitude and the phase being calculated from the response of the sensorsystem in a period in which the excitation generating unit has outputthe alternating current wave or square wave, by extracting a frequencycomponent of the alternating current wave or square wave from theresponse of the sensor system, and with vectors as Y and Z, which areobtained from an amplitude and a phase, the amplitude and the phasebeing respectively calculated from the response of the sensor system inperiods, immediately before and immediately after the aforementionedperiod, when the excitation generating unit does not output thealternating current wave or square wave, by extracting a frequencycomponent of the alternating current wave or square wave from theresponse of the sensor system, the demodulated signal corresponds to aconstant multiplication of |X−k·M| in which M represents a mean vectorof Y and Z, and k represents a coefficient whose value is determinedusing a response of the sensor system in a period when the excitationgenerating unit does not output the alternating current wave or squarewave.
 2. An electrostatic capacitance sensor including the electronicapparatus according to claim 1, comprising: a resistive sheet; and asensor system including a driving and detecting circuit connected to theresistive sheet for applying a voltage to the resistive sheet to measureand output current flowing through the resistive sheet, wherein atouching state or coordinates of an indicator are detected by detectingan electrostatic capacitance of a capacitor formed by the resistivesheet and the indicator.
 3. An electrostatic capacitance sensorincluding the electronic apparatus according to claim 1, comprising: anelectrode; and a sensor system including a driving and detecting circuitconnected to the electrode for applying a voltage to the electrode tomeasure and output current flowing through the electrode, wherein atouching state or coordinates of an indicator are detected by detectingan electrostatic capacitance of a capacitor formed by the electrode andthe indicator.
 4. An electrostatic capacitance sensor including theelectronic apparatus according to claim 1, comprising: a firstelectrode; a second electrode; and a sensor system including a drivingcircuit which applies a voltage to the first electrode and a detectingcircuit which measures and outputs current flowing through the secondelectrode, wherein a touching state or coordinates of an indicator aredetected by detecting an electrostatic capacitance of a capacitor formedby the first electrode and the second electrode.
 5. The electrostaticcapacitance sensor according to claim 2, comprising a display device,wherein a non-address period of the display device has a period when theexcitation generating unit outputs an alternating current wave or squarewave and a period when the unit does not output an alternating currentwave or square wave, and wherein the demodulated signal is generatedusing both a response of the sensor system in the period when analternating current wave or square wave is output and a response of thesensor system in the period when the alternating current wave or squarewave is not output.
 6. The electrostatic capacitance sensor according toclaim 3, comprising a display device, wherein a non-address period ofthe display device has a period when the excitation generating unitoutputs an alternating current wave or square wave and a period when theunit does not output an alternating current wave or square wave, andwherein the demodulated signal is generated using both a response of thesensor system in the period when an alternating current wave or squarewave is output and a response of the sensor system in the period whenthe alternating current wave or square wave is not output.
 7. Theelectrostatic capacitance sensor according to claim 4, comprising adisplay device, wherein a non-address period of the display device has aperiod when the excitation generating unit outputs an alternatingcurrent wave or square wave and a period when the unit does not outputan alternating current wave or square wave, and wherein the demodulatedsignal is generated using both a response of the sensor system in theperiod when an alternating current wave or square wave is output and aresponse of the sensor system in the period when the alternating currentwave or square wave is not output.
 8. The electrostatic capacitancesensor according to claim 2, wherein a value of the coefficient k isdetermined using both amplitudes: an amplitude of a forward noise as anoise obtained at the time of non-output of an alternating current waveor square wave before the excitation generating unit outputs analternating current wave or square wave; and an amplitude of a backwardnoise as a noise obtained at the time of non-output of an alternatingcurrent wave or square wave after the excitation generating unit outputsan alternating current wave or square wave.
 9. The electrostaticcapacitance sensor according to claim 3, wherein a value of thecoefficient k is determined using both amplitudes: an amplitude of aforward noise as a noise obtained at the time of non-output of analternating current wave or square wave before the excitation generatingunit outputs an alternating current wave or square wave; and anamplitude of a backward noise as a noise obtained at the time ofnon-output of an alternating current wave or square wave after theexcitation generating unit outputs an alternating current wave or squarewave.
 10. The electrostatic capacitance sensor according to claim 4wherein a value of the coefficient k is determined using bothamplitudes: an amplitude of a forward noise as a noise obtained at thetime of non-output of an alternating current wave or square wave beforethe excitation generating unit outputs an alternating current wave orsquare wave; and an amplitude of a backward noise as a noise obtained atthe time of non-output of an alternating current wave or square waveafter the excitation generating unit outputs an alternating current waveor square wave.
 11. A touch panel including the electronic apparatusaccording to claim 1, comprising: a resistive sheet; and a sensor systemincluding a driving and detecting circuit connected to the resistivesheet for applying a voltage to the resistive sheet to measure andoutput current flowing through the resistive sheet, wherein a touchingstate or coordinates of an indicator are detected by detecting anelectrostatic capacitance of a capacitor formed by the resistive sheetand the indicator.
 12. A touch panel including the electronic apparatusaccording to claim 1, comprising: an electrode; and a sensor systemincluding a driving and detecting circuit connected to the electrode forapplying a voltage to the electrode to measure and output currentflowing through the electrode, wherein a touching state or coordinatesof an indicator are detected by detecting an electrostatic capacitanceof a capacitor formed by the electrode and the indicator.
 13. A touchpanel including the electronic apparatus according to claim 1,comprising: a first electrode; a second electrode; and a sensor systemincluding a driving circuit which applies a voltage to the firstelectrode and a detecting circuit which measures and outputs currentflowing through the second electrode, wherein a touching state orcoordinates of an indicator are detected by detecting an electrostaticcapacitance of a capacitor formed by the first electrode and the secondelectrode.
 14. The electronic apparatus according to claim 1, comprisinga display device, wherein a non-address period of the display device hasa period when the excitation generating unit outputs an alternatingcurrent wave or square wave and a period when the unit does not outputan alternating current wave or square wave, and a demodulated signal isgenerated using both a response of the sensor system in the period whenthe alternating current wave or square wave is output and a response ofthe sensor system in the period when the alternating current wave orsquare wave is not output.
 15. The touch panel according to claim 11,comprising a display device, wherein a non-address period of the displaydevice has a period when the excitation generating unit outputs analternating current wave or square wave and a period when the unit doesnot output an alternating current wave or square wave, and a demodulatedsignal is generated using both a response of the sensor system in theperiod when the alternating current wave or square wave is output and aresponse of the sensor system in the period when the alternating currentwave or square wave is not output.
 16. The touch panel according toclaim 12, comprising a display device, wherein a non-address period ofthe display device has a period when the excitation generating unitoutputs an alternating current wave or square wave and a period when theunit does not output an alternating current wave or square wave, and ademodulated signal is generated using both a response of the sensorsystem in the period when the alternating current wave or square wave isoutput and a response of the sensor system in the period when thealternating current wave or square wave is not output.
 17. The touchpanel according to claim 13, comprising a display device, wherein anon-address period of the display device has a period when theexcitation generating unit outputs an alternating current wave or squarewave and a period when the unit does not output an alternating currentwave or square wave, and a demodulated signal is generated using both aresponse of the sensor system in the period when the alternating currentwave or square wave is output and a response of the sensor system in theperiod when the alternating current wave or square wave is not output.